EEWeb Pulse - Issue 62

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INTERVIEW EEWeb.com Issue 62 September 4, 2012

Sailesh Chittipeddi President and CEO Conexant Systems

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TABLE OF CONTENTS

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Sailesh Chittipeddi CONEXANT SYSTEMS Interview with Sailesh Chittipeddi - President and CEO

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Featured Products Conexant’s Far-Field Voice Input Processing

BY SAILESH CHITTIPEDDI WITH CONEXANT

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Conexant, market leader in audio with over 300M units shipped over 5 years, enables true far field voice input processing and speech recognition capabilities with its latest Skype certified offerings.

Hidden Hazards in the Sallen-Key 2nd Order High Pass Active Filter

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BY MICHAEL STEFFES WITH INTERSIL Simple ways to improve performance and fix flatness in the desired signal passband.

Microampere Current-Sense Amplifiers: Redefining a New State-of-the-Art

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BY ADOLFO GARCIA WITH TOUCHSTONE How new CSA enhancements enable the next generation of battery-powered, hand-held portable instruments addressing power management, motor control, and fixed-platform applications.

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RTZ - Return to Zero Comic

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Sailesh

Chittipeddi Conexant Conexant Systems is a leading provider of solutions for imaging, audio, embedded modem and video surveillance applications based in Newport Beach, California. We spoke with the President and CEO, Sailesh Chittipeddi about what it means to be in the “true” far-field arena, the company’s rich heritage of technology and IP and their plans for long-term growth.

How did you begin your engineering career?

I started my career at Bell Laboratories in Allentown, Pennsylvania with the CMOS technology development group. I spent quite a bit of time working on process technologies development and then on transferring those processes into manufacturing.

When companies owned their wafer fabrication facilities, as was the case with AT&T Microelectronics and subsequently Lucent Technologies Microelectronics

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Division, the linkage between design engineering and process technology development was very tightly coupled to drive specific customer requirements. During this stage in my career, I had an opportunity to work with some of the best and brightest engineers from a process technology as well as design perspective. As a result of this activity when I was with Bell Labs, I was issued 63 U.S. patents and wrote about 40 publications. I transitioned sometime in the late 90s into the operations side of the business.

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In 2001, I took over the wafer foundry management responsibility for Lucent Technologies Microelectronics Division as we started the transition to a fabless model.

Lucent Technologies Microelectronics Division was later spun-off to become Agere Systems.

I began working very closely with foundries, transferring our process technologies to the outside. In 2006, I moved over to Conexant Systems and became senior vice president of global operations. Subsequently, I assumed responsibility for the

INTERVIEW central engineering efforts and took on the role of CTO for a period of time, after which I became the president and COO prior to assuming my current role as CEO. What are the main areas of technology that your company provides?

Our technology focus and new investments are in two primary areas – audio and video. One of the areas of focus in the audio business is the smart television market with voice and speech recognition capabilities.

We bring our far-field voice input processing knowledge into this market space via our voice input processor SoCs. One of the latest televisions demonstrated by Korean manufacturers uses our voice input processor SoC, and it is currently selling into the market today. In terms of algorithm development, far field voice input processing is where we have a niche. We are in the true far field arena, which means if you are sitting in a room and are watching television and you tell the television to wake up, it basically wakes up. We are currently developing a new technology called “Watch, Live and Talk.” Imagine you have your television at full-blast and are watching a baseball game when you get a Skype conference call from the other side. This technology will allow you to take the call and watch the game at the same time without the person on the other side hearing what you are watching.

The second area in which we are involved in audio is the unified communication headset market.

We are partnering with every major headset manufacturer and have a dominant presence in that market.

The third area is what we broadly call integrated mobile audio, which includes two specific facets: PC audio and mobile audio. Our

analog and mixed signal design together with our firmware enables us to provide customers with a differentiated audio and voiceinput processing experience. It is what distinguishes us from the competition.

Video is a significantly smaller business for us, but we expect it to scale up significantly from current levels over the next several years,

If you step back for a moment and ask how Conexant is different from all of the broad-based audio suppliers, you’ll see that we don’t compete on an individual, commoditized component level and take advantage of the move from standard-definition to high-definition technologies with our new product launches. In video, the application segments in which we are focused include security and monitoring and video surveillance. We also have a legacy PC-TV business that is being adapted for a broader application base re-using existing IP. The application processor used for security and monitoring markets can also be adapted for low-cost tablets and is currently being used

by Datawind and other suppliers to address emerging market needs.

In addition to these two investment areas, we still have a fairly mature fax and embedded modem business, where we continue our marketleading positions. Unlike some of our competitors, we are investing in roadmap development to sustain our leadership positions and make sure our customers’ needs are met.

Unlike the PC-modem, which is largely de-bundled, the continual demand for secure point-to-point communications keeps the demand for these businesses relatively healthy.

The final product line where we had made substantial investments in the past is the Imaging Systems Group, which focuses primarily on SoCs for multi-function and single-function printers. This unit has successfully deployed 11 generations of printer SoCs. The combination of its unique hardwired imaging pipeline combined with superior firmware development makes it a market leader in the ASSP market for printer SoCs. Between your audio business and video business, which would you say is more dominant in terms of direction of resources you plan to hit?

Audio is a very resource intensive area for us. We are selectively investing in video, but not in the conventional areas where we would run into major SoC participants.

Rather, we are focused in areas where we see more opportunities to bring our competencies in design and software/firmware to provide differentiation. So when you look at it in aggregate, audio R&D investment is very heavy – we are invested in a wider range of applications than on the video side, but in the next 18- 24 months, we expect video to contribute to a significant portion of our growth.

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...our far field processing algorithms allow us to accomplish with one omnidirectional microphone what most of the competition are doing with two to four.

How has Conexant become a leading power of solutions for imaging, audio, embedded modem, and video surveillance applications?

In the imaging area, which is a combination of fax and printer SoCs, traditionally we have had fax datapumps and fax SoCs in our pipeline, and the entry barriers to newcomers in that market is fairly high. In 2008, we acquired the Sigmatel (Oasis) Printer SoC business from Freescale and have invested to make it successful.

This investment has also led to our recent successes in other areas such as the low-cost tablet business in India. In embedded modems, we have a mature business, which is expanding into the ePOS segment in China. Our significant R&D investment moving forward is primarily in audio and video areas.

Specifically, our analog-mixed signal capability, coupled with what I would characterize as general signal processing knowledge and firmware/software capabilities, allows us to pursue niches that large players would typically not pursue given the level of support required.

In the algorithm development area,

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for example, we have differentiated technologies in Subband AEC, stereo AEC, Phantom Bass and Brightsound XV- playback processing.

Conexant sell ICs or is it primarily cores for ASICs? We sell ICs, but we don’t just sell silicon – the ICs will have a firmware component to it. We won’t compete in a pound-by-pound basis with somebody else, so almost anything that we do will have some differentiated analog and mixed signal capabilities with a software and firmware element. If you step back for a moment and ask how Conexant is different from all of the broad-based audio suppliers, you’ll see that we don’t compete on an individual, commoditized component level – rather we look for something that will have a component plus a software or firmware play that gives our customer a differentiated advantage in the end-markets in which they compete.

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Is Conexant a fabless company?

Yes. Conexant was spun out of Rockwell Systems Semiconductor Group in 1999. Since that point, we’ve gone through a bunch of acquisitions like Globespan in 2003 and have spun off businesses like Jazz Semiconductor, Skyworks Solutions, Mindspeed Technologies, SiRF and Pictos. There is a lot of rich IP in this company, so the analogy to Bell Labs and AT&T Microelectronics/ Lucent Technologies/Agere Systems is quite telling because both companies have such a rich heritage in technology. If one looks back at it, the challenge for both organizations was the inability to capitalize on the tremendous knowledge and IP resident in those organizations. We are now focused on markets and applications where we have the engineering and core competencies to enable us to be successful and resume our growth.

The advantage we currently have since going private in 2011 is that we can afford to take a longer-range

INTERVIEW view of our investments and take the necessary steps consistent with that view to be successful.

How involved are your technical resources with the customer? Does Conexant have a consulting service integrated with the product?

Every headset or television opportunity will involve the engineers working very closely with the customer. The reason is pretty simple – every television Webcam (external or embedded) has a different enclosure design, so, forexample, the characteristics that are associated with processing are going to be very different based on how flat or thick your television is and on exact speaker placement. The same applies for headsets. We have to work very closely with customers to tune the algorithms that will make them successful to give them the experience they desire. One thing that makes Conexant unique is that we are the only certified omni directional microphone Skype solution on the market today. Before, you would have solutions requiring a lot of microphones, but our far field processing algorithms allow us to accomplish with one omnidirectional microphone what most of the competition are doing with two to four uni-directional microphones.

What are some challenges you have faced since being named president and CEO last year?

The company has a rich heritage of technology and IP in its portfolio.

However, financially the debt overhang from its early existence as a public company, coupled with a global real-estate footprint of a multi-billion dollar company, has left the company in a continual state of re-engineering without a

focused investment strategy. What we have started doing in the last 18 months, and continue to do more of (since going private has given us the flexibility to do so), is focus our investments in a few select areas, get our financial house in better shape, and execute much better both from an engineering and operational perspective than we ever have done before. This has allowed us to focus on market niches that are very suitable to attack for a company of our size and scale.

What challenges do you foresee in the industry?

The slow-growth in the worldwide economy may put a crimp on overall industry growth in the near term, in the event the economic powers that be do not resolve the mounting debt crisis enveloping most economies.

The migration from PCs to tablets has reduced semiconductor content affecting traditional DRAM, as well as more traditional wired component players, but it has fueled the growth for other applications. Longerterm, the increased applicability of semiconductors for a variety of consumer applications ranging from home-entertainment, smart-home, medical, security, surveillance, connectivity (be it mobile or TV), sensors and a variety of industrial and automotive applications will drive growth of the industry as well as for Conexant. Additionally, the need for increased storage (both from an enterprise and consumer perspective), as well as the continual demands for increased bandwidth and ubiquitous connectivity, will ensure the semiconductor industry growth over the longer-term

company considerably. One of the advantages we have by remaining private is that we can take a longer term view versus a public company.

In addition to organic growth, we continually assess acquisition opportunities that will either provide us with the IP or allow for a portfolio expansion. While there is pressure to deliver results, we are focused more on medium/long-term growth objectives, as opposed to shortterm quarter to quarter targets.

How many employees are there at Conexant? How would you describe the culture within the company?

There are about 450 people currently working at Conexant. The word I used to describe the culture in the new Conexant is resilience. I would say the company’s culture is changing considerably, so our staff is certainly adaptable to change, and continues to have a desire to be a learning organization. The employees and engineering teams have been through a lot, and there is a tremendous desire to be successful and turn the place around. That’s what keeps me here – the fact that the team truly wants to succeed. There are a lot of competing companies close by, yet our extremely talented staff chooses to stay with us, because they want to move the company forward. ■

Can you tell us about some other goals for Conexant?

Our main objective for the next five years would be to grow the Visit www.eeweb.com

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Conexant’s Far-Field Voice Input Processing by

Sailesh Chittipeddi

Think of the advantages of carrying on a telephone

conversation without having to worry about where the microphone is – simply talk and some hidden technology will pick up your voice, filter out any extraneous noise and send it to the other side. Just like talking to someone in the same room. No holding a phone to your ear, no headset to wear, no microphone to stay close to. And suppose we were able to control the gadgets around us simply by talking to them.

No buttons to push or remote controls to find – we simply tell them to do what we want. Truly hands-free operation.

Past attempts at providing this functionality have largely failed because of the difficulty with what is termed ‘far-field voice input processing’ (FFVIP).

In near-field voice input processing, where a microphone is close to the mouth of a person talking, the audio quality tends to be quite good and louder than surrounding noise. When there are disturbances, several techniques are available to distinguish the

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near-field talker from far-away disturbances. Such techniques include using spatial differences from multiple microphones, as well as taking advantage of the level difference to distinguish the higher-level voice from the noise, then combining that with statistical algorithms to deliver a high-quality voice signal output.

However, if the microphone is 12-15ft (4-5m) away, then the voice signal is part of the far-field and it can be buried in a variety of disturbances – for example, echo from nearby loudspeakers, noise from traffic or appliances, reverberation from the walls of the room, and even other voices nearby. In these cases, the voice level can be much lower than the noise sources, and in the case of sound from a loudspeaker located in the same unit as the microphone (e.g. a speakerphone) that echo can easily be 100 times louder than the desired voice signal.

To deliver a clear, easily understandable voice signal from a far-field source requires a new set of advanced algorithms. Conexant’s FFVIP technology addresses the following:

PROJECT To suppress the echo from a local loudspeaker by a factor of a million is extremely challenging. To address this, Conexant has developed algorithms that use advanced adaptive filters to estimate the echo and perform statistical estimation of which frequency bands contain echo and which contain the desired voice. Sophisticated control algorithms tie these together to produce a natural-sounding echo-free voice signal.

Background Noise Suppression

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When trying to provide true FFVIP, there are a variety

of background noises that must be eliminated. Not just stationary noise from fans and motors but also time-varying noise from passing cars, airplanes, home appliances and office machines. To eliminate background noise, Conexant has developed algorithms that analyze the spectral and temporal characteristics of the microphone signal and suppress anything that is clearly not voice without impacting real voice signals.

De-Reverberation When speaking inside a room, the voice signal bounces off walls and often takes hundreds of milliseconds to die down. Normally the human auditory

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Echo Cancellation

Near Field Experience Far Field Experience Visit www.eeweb.com

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Stationary or Spatially Coherent

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system sorts these reverberations out so they’re not noticeable. However, if the sound is recorded by a microphone it becomes very pronounced and sounds as if the person is speaking from a large empty barrel. This reverberation is not only disturbing, but affects speech comprehension both for humans and speech recognition engines. Conexant’s unique de-reverberation algorithm solves the challenge of removing the reverberation without adding other artifacts.

Gain Level Adjustment To keep the voice signal at a constant level independent of the distance to the microphone, the gain level needs to be adjusted without changing the voice signal characteristics. Specifically, low-level sounds need to stay low and high level sounds need to keep their

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emphasis. Keeping the voice level too constant will make it sound unnatural and it will lose some of its characteristics.

With its collection of audio-processing algorithms that enable devices to pick up a clear voice signal, even if it’s generated far away in a noisy background, Conexant’s FFVIP technology removes a major stumbling block for the integration of accurate voice control and speech recognition capabilities in today’s leading consumer electronic devices.

FFVIP enabled devices can pick up a clear voice signal from a noisy background. This results in good hit-rate for the speech recognition algorithms and reliable voice control from far away. Without FFVIP, voice commands have to be issued very close to the appliance, which limits their use significantly.

PROJECT For example, this year’s high-end smart TVs come enabled with two significant new features: they support accurate voice recognition and they allow the user to make reliable Skype TV calls. Both of these exciting features are enabled by Conexant’s revolutionary FFVIP technology.

The effectiveness of Conexant’s FFVIP technology has been recognized by many leading consumer application manufactures, which have started to implement this technology in a wide array of consumer products such as smart TVs, notebooks, home appliances and more. Some of these products are already available in the market and others will be available in the near future.

References 1.)http://eon.businesswire.com/news/ eon/20120208005291/en/Conexant/CX20805/DigitalAudio-Processor 2 . ) h t t p : / / w w w. b u s i n e s s w i r e . c o m / n e w s / home/20110525005343/en/Conexant-Delivers-SuperWideband-Audio-Chip-Home 3.)http://developer.skype.com/certification/odmprogram/adc-voice-processing-soc-1-omni-mic 4.)http://developer.skype.com/certification/odmprogram/adc-beamforming-soc-2-mic-array

Conexant’s FFVIP technology can be made available in three solutions based on customer needs. The first option is for Conexant’s CX 208051 high-performance audio DSP with integrated power management and CX207082 voice input processor to be used as standalone products.

The second option involves hardware IP on Conexant’s DSP core, in which the FFVIP algorithms have been ported to Conexant Audio Processing Engine (CAPE), a hardware IP core available for integration in SOC designs. The CAPE architecture supports the most efficient compilation of DSP code written in pure C, with minimal use of macros and intrinsics.

The third option involves software IP, in which the algorithms are also available as a software IP package that can be ported to the customer’s SOC. Conexant Hardware voice input processor and algorithms are certified by Skype in both single mic3 and dual mic4 versions.

Conclusion The challenge with far-field voice input processing lies in bringing multiple algorithms to bear on voice signals to remove distortion and still provide a clean and natural-sounding voice signal. Using its audio digital signal processors, Conexant has developed a set of advanced algorithms that remove the noise, echo and reverberation, and enhance the voice signal to deliver high-quality voice through far-field pickup to make true hands-free operation a reality.

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TECH ARTICLE

dden azards

e Sallen-Key Order High s Active Filter

chael Steffes

sil - Sr. Applications Manager

The simple Sallen-Key Filter (SKF) applied to high pass requirements appears fairly straightforward and simple to implement.

Latent within the design is a risk of capacitive and/or heavy loading for the op amp driving into this active filter stage. Most developments assume an ideal voltage source for the input signal where the effect of a reactive or heavy load would be hidden. When multistage designs are intended using low power op amps, the load presented by a poorly designed stage might in fact impair the intended signal frequency response above the high pass corner. Some of those tradeoffs will be exposed here with example designs. Simple paths to improved performance will be shown where improved flatness in the desired signal passband can be achieved.

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R1

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Figure 1: 2nd order, SKF high pass filter

The SKF High Pass Filter

Contrary to the SKF low pass design, the area of interest for the signal is actually above the high pass corner frequency of the filter. Looking at figure 1, as the frequency increases above Fo, the caps short out and the source simply sees R2 terminating into a noninverting gain stage. But what about the effect of the R1 path at frequencies above the intended high pass cutoff? Considering a gain of 1 design, the same signal that appears at the input to R1 in the passband appears at the output of the amplifier -effectively bootstrapping out this path from changing the input impedance away from just R2. As the frequency continues to increase, the propagation delay and rolloff through the amplifier will cause R1 to appear as more of a load in parallel with R2.

It is in fact very possible that this active impedance path can dominate the total input impedance presenting a load much lower than just the R2 element. Some design points are worse than others in this respect.

The classic Sallen-Key Filter (SKF), also known as a voltage controlled voltage source filter ( VCVS), design Here, the design will begin by constraining R2 to min/ for a 2nd order high pass is shown in Figure 1. This is max ranges. Increasing R2 will help the loading on the prior stage but comes at the cost of the higher noise following the numbering of ref. 1, page 399. contribution around Fo and possibly added input Here, the amplifier acts to convert a passive RC network offset consuming signal headroom by the op amp bias into a design that can offer complex poles in the current times this resistor. One aspect of this design is to implementation of a high pass filter. The ideal transfer balance this R2 issue with the resulting R1 to hit the filter function for this network, where K = 1+Rf/Rg, is given shape which then also comes into the input impedance characteristic. To pursue this, the link between R2 and as Eq. 1.

R1 over different design choices must be developed.

Ks2 V out

= V in s2 + s R12 C11 + C12 +

1 R1 R2 C1 C2

The Resistor Ratio Constraint in the SKF High Pass Design.

The amplifier gain provides the higher frequency gain setting for the signal path and is part of the Q setting equation. Either Voltage Feedback Amplifier (VFA) or Current Feedback Amplifier (CFA) type devices can be used in this circuit.

All SKF filters achieve their Q as a combination of amplifier gain (K), capacitor ratio and then resistor ratio.

If one of our design goals is to keep R1 from getting too low, what might create that condition? If the capacitor ratio is swept for a particular amplifier gain and desired Q, the required R2/R1 ratio can be generated using the relationship of Eq. 4 (ref.2).

1−K R1 C1

+

The characteristic frequency and 1/Q for this transfer function is given in equations 2 and 3.

ω0 =

1 = Q

18

R2 R1

C1 + C2

1 R1 R2 C1 C2

C2 C1

− (K − 1)

2Q (1 + α)

β=√

2 α 1 + 1 + 4Q (1 + α) (k − 1)

Here, K is the amplifier gain, Q is the target for the filter poles, with

R1 C1 R2 C2

EEWeb | Electrical Engineering Community

α=

C2 C1

and

β=

R2 R1

Most design references assume the gain of 1 design is most advantageous for amplifier bandwidth and sensitivity reasons. However, it turns out this conditions always requires that R1<R2 (β>1) – and sometimes significantly less. Sweeping the C ratio from about 0.2 to 5 for a gain of 1 and plotting the R2/R1 ratio for different target Q’s gives Figure 8. It is desirable that this ratio be low. Using equal C gives the minimum but as the required Qp goes up, R1 must be much lower than R2 as shown in the log/log plot of Figure 8.

R2/R1 Ratio vs C2/C1 Ratio K=1 Q=.577

Q=.707

Q=1

Q=5.27

R2/R1 Ratio

1000

100

TECH ARTICLE

Example design for 1kHz 2nd order high pass with >1Mhz signal bandwidth for K=1 The lowest R2/R1 ratio in figure 2 is for equal C at K=1.

Use the ISL28113 (ref.3) to get a signal bandwidth exceeding 1Mhz in a μPower design as shown in the circuit of figure 4 (ref.4). This device offers a 2MHz Gain Bandwidth Product (GBP) using only 90μA (typical, 130μA max) supply current on a 1.8V to 5.5V supply.

Use an R2 that adds an input noise approximately equal to the amplifier’s 25nV/√Hz and start the design with R2=50kΩ. This high Q design will be peaking the input noise around Fo quite a lot, so it is best to not let R2 get too high. The design of Figure 4 used 50k for R2, but that forced R1 down to 457Ω using Eq. 4.

457 R1

10 33.3n

20

C2

Rb

33.3n

Vin

1 0.1

1

10

C1

R2

50k

ISL28113 + –

C2/C1 Ratio

Figure 8: Required R2/R1 ratio for swept C2/C1 parametric on Q using K=1.

Getting some of the Q with gain in the amplifier has a dramatic effect on this in a desirable direction.

Using an amplifier gain of 2 and repeating this same calculation gives the required R2/R1 ratio of Figure 9.

Using just a bit of gain has moved the required R1 value up significantly if R2 is chosen for loading, noise, and input offset voltage reasons. These curves also suggest selecting C2/C1>1 might be desirable.

R2/R1 Ratio vs C2/C1 Ratio K=2 Q=.577

Q=.707

Q=1

Q=5.27

R2/R1 Ratio

10

1

0.1 0.1

1

10

C2/C1 Ratio

Figure 9: Required R2/R1 ratio for swept C2/C1 parametric on Q using K=2.

Using these two gains of 1 and 2, the difference in input impedance will be shown for a Q = 5.27 design (this is the approximate highest Q stage required for a 6th order 0.25dB Chebychev filter).

V+

+

1.25u

–

V– 10k Rf

-9.29209u

Figure 10: High Q, K=1 design with 1kHz Fo and Q = 5.27

Figure 11 shows the expected frequency response while Figure 12 shows the relatively high noise peaking around Fo. The response curve is showing the expected peaking at 1kHz, and then a gain of 1 over a broad passband with the amplifier rolling off above 2Mhz. The output spot noise peaks approximately 60X around Fo. This is common for high Q stages but is even higher here due to the very high resistor ratio. Since this is happening at lower frequencies it should not impact the integrated noise too much, but will be degrading the loop gain at the lower end of the intended passband.

Reducing this noise gain peaking for high Q poles is desirable and easily achieved by adding some gain in the amplifier.

The added concern in Figure 10 is the several regions of capacitive input impedance. Figure 13 shows simulated input impedance showing the initial capacitive response up to Fo which then recovers to the R2 resistor value.

The phase response across the R1 resistor comes down to approximately 0deg above Fo over a wide frequency range effectively bootstrapping out the relatively low R1 value. However, even a slight phase deviation over the Visit www.eeweb.com

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dbV@VOUT/dB

10 0 -10 -20 -30 -40

100

200

400

1k

2k

4k

10k

20k

40k

100k 200k 400k

1M

2M

4M

10M

Frequency/Hertz

10u 4u 2u

Running a frequency response and probing the output of the first stage (red curve) along with the final output gives the desired high pass complex pole filter shape but now adds perhaps an undesirable peaking at higher frequencies.

1u

10

400n

0

200n 100n

-10

40n

dB

Output Noise / V/rtHz

Figure 11: Gain of 1, 2nd order SKF high pass frequency response.

impedance profile impacts the overall performance depends strongly on the specific device driving into this load. Using an ISL28136 (ref. 5) will give the circuit of Figure 14. This slightly lower noise device is faster and hence more susceptible to capacitive load peaking issues. This is shown as just a buffer stage here, but normally this would be another active filter stage to implement a multipole high pass filter.

20n

-30 100

200

400

1k

2k

4k

10k

20k

40k

100k 200k 400k

1M

2M

4M

10M

-40

Frequency/Hertz

-50

Figure 12: Gain of 1, 2nd order SKF high pass output spot noise

100

400

1k

2k

4k

10k

20k

40k

100k 200k 400k

1M

2M

4M

10M

Figure 15: Buffered 2nd order SKF HPF response showing input impedance effects.

40k 20k

VIN / V

200

Frequency/Hertz

100k

10k 4k 2k 1k 400 200 100

200

400

1k

2k

4k

10k

20k

40k

100k 200k 400k

1M

2M

4M

10M

Frequency/Hertz

Figure 13: Input impedance to the design of Figure 10.

intended signal frequency span causes the apparent input impedance to vary widely and extend to very low values as shown in Figure 13.

This impedance looks roughly like a capacitance again above 20kHz. If this stage is then driven from the output of another amplifier stage, some impact on the response flatness of that device should be expected. Whether this 457 R1

V+ X1

Vin

+

ISL28136

+

– –

V–

33.3n

20

C2

Rb

33.3n C1

R2

1.25u

50k

ISL28113 + –

V+

+

Vout –

V– 10k Rf

-9.29209u

Figure 14: High Q, K=1 HPF driven by another amplifier stage.

20

-20

This issue was showing up in the construction of an online semi-automatic multi-stage high pass active filter design tool. Many amplifier and impedance combinations are possible, but using a gain of 1 for the higher Q stages introduces a very wide component ratio spread that causes other problems as well. While increasing the gain for the highest Q stage seems like it is going in the wrong direction, it is actually possible numerous 2nd order benefits will be seen in physical implementations.

Using K=2 in the SKF HPF to improve the input impedance characteristic.

The parametric R vs. C ratio curves shows a significant reduction in required R ratio with that addition of some gain in the amplifier. Then, starting from an R2 value that does not impact the total noise too much, using a K=2 will pull up the required R1 value nicely. Continuing with the equal C design for simplicity and holding R2 = 50kΩ adjusts the C values down and the R1 value up for 1kHz, Q = 5.27 design as shown in figure 9. With R1 resolved to 28.6kΩ using eq. 4, the R1C product will be given by eq. 5 for k>1 (letting k=1 gives the solution for the C in the first example). Dividing this result by R1, gives the value for the equal C in this design flow.

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1 2 R1 C = 1 + 1 + 8Q (k − 1) 4ω0 Q 28.6k R1 4.2n

20

4.2n C2

C1

R2

50k

V+

+

+

Most interesting is the significant change in input impedances between the two designs. Getting the R1 value up closer to the R2 value causes less capacitive characteristic at high frequencies along with higher minimum impedance as shown in the green curve of Figure 20.

Vout

Rb –

–

1M

V– 10k

1.25u R4

10k

100k

Rf

VIN / V

Vin

ISL28113

TECH ARTICLE

-18.8333u

Figure 17: K=2 design for Fo = 1kHz, Q = 5.23, low power HPF using the ISL28113.

10k

1k

The response shows the same high pass poles as the unity gain design (shifted up 6dB) but of course lower high end cutoff frequency. The frequency response for the unity gain design and this K=2 design are shown in Figure 18. These responses are not following a strict gain bandwidth product profile due the correctly modeled open loop phase effects in the ISL28113 macromodel. It is not uncommon to see a bit of bandwidth extension in VFA based designs operating at lower gains where the phase margin is <70 degrees.

200

100

400

1k

2k

4k

10k

100k 200k 400k

1M

4M

10M

Now adding the ISL28136 as a buffer stage into this design gives the circuit of Figure 21.

28.6k R1

V+ X1

Vin

+

ISL28136

+

4.2n

20

C2

Rb

4.2n

–

C1

–

R2

50k

ISL28113 + –

V+

+

Vout –

V– 10k

1.25u

10

R4

0

10k

Rf

-10

-18.8333u

Figure 21: Buffered input to the K=2 design.

-30 -40 200

400

1k

2k

4k

10k

20k

40k

100k 200k 400k

1M

2M

4M

10M

Frequency/Hertz

Figure 18: Response shape comparison for gain of 1 (black) and gain of 2 single stage designs.

Implementing the poles with a bit of amplifier gain will also change the output noise profile as shown in Figure 19. While the noise gain peaking around Fo =1kHz has been reduced in this new design (red curve) but the wider band span runs at higher noise due the K=2 setting.

10u 4u 2u 1u 400n 200n

Looking at the frequency response for this implementation at both the output of the ISL28136 buffer and the final filter output gives the considerably improved high frequency flatness of Figure 22.

20 0 -20

dB

-20

100

2M

Figure 20: Improved input impedance for K=2 design (black curve) vs. K=1.

V–

dB @ VOUT/dB

40k

Frequency/Hertz

20

Output Noise / V/rtHz

20k

-40 -60 -80

100

200

400

1k

2k

4k

10k

20k

40k

100k 200k 400k

1M

2M

4M

10M

Frequency/Hertz

Figure 22: Frequency response for the buffer output (red curve) and total response to Vout.

100n

The design at K=2 has still achieved the desired high Q high pass poles, but now shows a much more controlled 100 200 400 1k 2k 4k 10k 20k 40k 100k 200k 400k 1M 2M 4M 10M high frequency response. Operating at a K=2 pulled Frequency/Hertz the R1 element up closer to R2 giving a more benign Figure 19: Spot output noise comparison between gain of input impedance over frequency.

40n 20n

1 and 2 designs.

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Summary and Conclusions

About the Author

Gain of 1 in the amplifier for a high pass SKF has often been the preferred approach in the design and vendor literature. Using real amplifiers, or macromodels that show the impact of load impedance on response shape, can run into response flatness issues driving into the reactive load impedance seen at frequencies > Fo. Assuming the region of signal interest is actually above the high pass corner, this peaking might be totally unacceptable in a physical implementation. One path to shifting this loading issue in a good direction might be to take advantage of the amplifier gain available in the design to pull the R ratio much closer. This has been shown here to be an effective means of improving the response flatness. This issue is very dependent on the specific amplifiers chosen for the design but simulation tools are readily available to the designer to easily evaluate options (ref. 4). If your multi-stage HP SKF design has been showing response peaking above the high pass corner, perhaps it is this loading issue internal to the filter and only a slight design change to improve the input impedances can quickly improve your response flatness.

With 27 years of involvement in high speed amplifier design, applications, and marketing, Michael Steffes has introduced over 80 products spanning five companies while publishing more than 40 technical articles. His current focus is on high efficiency high speed ADC interfaces, DSL/PLC line interface solutions, and online design tool development. ■

References 1. “Passive and Active Network Analysis and Synthesis”, Dr. Aram Budak, 1974, pp 399 2. This is essentially the same equation as developed for the SKF Low pass with the definition of α and β reversed and then each ratio reversed. Contact the author for the detailed derivation for the SKF low pass version.

3. ISL28113, Single General Purpose Micropower, RRIO Operational Amplifier, http://www.intersil.com/ content/intersil/en/products/amplifiers-and-buffers/allamplifiers/amplifiers/ISL28113.html 4. These circuits (available from the author) come from the free Spice simulator (registration required), iSim PE available at http://www.intersil.com/en/tools/isim.html 5. ISL28136, 5MHz, Single Precision RRIO Op Amp, http://www.intersil.com/content/intersil/en/products/ amplifiers-and-buffers/all-amplifiers/amplifiers/ ISL28136.html

22

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TECH ARTICLE Get the Datasheet and Order Samples http://www.intersil.com

Single or Multiple Cell Li-ion Battery Powered 4-Channel and 6-Channel LED Drivers ISL97692, ISL97693, ISL97694A The ISL97692, ISL97693, ISL97694A are Intersil’s highly integrated 4- and 6-channel LED drivers for display backlighting . These parts maximize battery life by featuring only 1mA quiescent current, and by operating down to 2.4V input voltage, with no need for higher voltage supplies.

The ISL97692 has 4 channels and provides 8-bit PWM dimming with adjustable dimming frequency up to 30kHz. The ISL97693 has 6 channels with Direct PWM dimming control.

The ISL97694A has 6 channels and provides 8-, 10-, or 12-bit PWM dimming with adjustable dimming frequency up to 30kHz, 7.5kHz, or 1.875kHz, respectively, controlled with I2C or PWM input.

ISL97692 and ISL97694A feature phase shifting that may be enabled optionally, with a phase delay between channels optimized for the number of active channels. In ISL97694A, phase shifting can multiply the effective dimming frequency by 6 allowing above-audio PWM dimming with 10-bit dimming resolution.

The ISL97692/3/4A employ adaptive boost architecture, which keeps the headroom voltage as low as possible to maximize battery life while allowing ultra low dimming duty cycle as low as 0.005% at 100Hz dimming frequency in Direct PWM mode.

The ISL97692/3/4A incorporate extensive protection functions including string open and short circuit detections, OVP, and OTP. The ISL97692/3 are offered in the 16 Ld 3mmx3mm TQFN package and ISL97694A is offered in the 20 Ld 3mmx4mm TQFN package. All parts operate in ambient temperature range of -40°C to +85°C.

Features • 2.4V Minimum Input Voltage, No Need for Higher Voltage Supplies • 4 Channels, up to 40mA Each (ISL97692) or 6 Channels, up to 30mA Each (ISL97693/4A) • 90% Efficient at 6P5S, 3.7V and 20mA (ISL97693/4A) • Low 0.8mA Quiescent Current • PWM Dimming Control with Internally Generated Clock - 8-bit Resolution with Adjustable Dimming Frequency up to 30kHz (ISL97692/4A) - 12-bit Resolution with Adjustable Dimming Frequency up to 1.875kHz (ISL97694A) - Optional Automatic Channel Phase Shift (ISL97692/4A) - Linear Dimming from 0.025%~100% up to 5kHz or 0.4%~100% up to 30kHz (ISL97692/4A) • Direct PWM Dimming with 0.005% Minimum Duty Cycle at 100Hz • ±2.5% Output Current Matching • Adjustable Switching Frequency from 400kHz to 1.5MHz

Applications • Tablet, Notebook PC and Smart Phone Displays LED Backlighting

Related Literature (Coming Soon) • AN1733 “ISL97694A Evaluation Board User Guide” • AN1734 “ISL97693 Evaluation Board User Guide” • AN1735 “ISL97692 Evaluation Board User Guide” 10

L1

VIN: 2.4V~5.5V

10µH

4.7µF

10

D1

VOUT: 24.5V, 6 x 20mA 4.7µF 4.7µF

1

VIN LX COMP

15nF 12k

100pF 470k

OVP 2.2nF

ISL97694A

ILED (mA)

1µF

23.7k

ISET 53k

AGND

PGND

0.1

0.01 fPWM: 200Hz

SDA/PWMI SCL

CH1

EN

CH2

FPWM 291k

143k

FSW

0.001

CH3 CH4 CH5

0.0001 0.001

CH6

FIGURE 1. ISL97694A TYPICAL APPLICATION DIAGRAM

July 19, 2012 FN7839.2

fPWM: 100Hz

0.01 0.1 1 INPUT DIMMING DUTY CYCLE (%)

10

FIGURE 2. ULTRA LOW PWM DIMMING LINEARITY

Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2012 All Rights Reserved. All other trademarks mentioned are the property of their respective owners.

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TECH ARTICLE

High-Side Current-Sense Amplifiers:

Delivering Greater Performance Adolfo Garcia

Touchstone Semiconductor Vice President – Marketing & Applications

To sense and control supply current flow are fundamental requirements in most electronic systems from battery-operated, portable equipment to mobile or fixed-platform power management and dc motor control. High-side current-sense amplifiers (or “CSAs”) are useful in these applications especially where power consumption is an important design parameter. In allowing engineers to save power without sacrificing performance, a new breed of CSAs offers even greater benefits.

Design engineers now have even more options for high-side current-sensing amplification with the right combination of wide operating supply-voltage range, low supply-current operation, low input offset voltage (VOS) and gain errors, fixed gain options, and small form factors. Addressing power management, motor control, and fixed-platform applications, new CSA enhancements now enable the next generation of battery-powered, hand-held portable instruments.

Uni-directional Current-Sense Amplifiers For measuring load currents in the presence of highcommon-mode voltages, the internal configuration of some uni-directional CSAs is based on a commonly-used operational amplifier (op amp) circuit. In the general case, a CSA monitors the voltage across an external sense and generates an output voltage as a function of load current. Featuring Touchstone Semiconductor’s TS1100, the inputs of the op-amp-based circuit are connected across an external RSENSE as shown in the typical application circuit in Figure 1. The applied voltage is ILOAD x RSENSE at the RS- terminal.

Op-amp feedback action forces the inverting input of the internal op amp to the same potential (ILOAD x RSENSE) since the RS- terminal is the non-inverting input of the internal op amp. Therefore, the voltage drop across RSENSE (VSENSE) and the voltage drop across RGAIN (at the RS+ terminal) are equal. Both RGAIN resistors are the same value to minimize any additional Visit www.eeweb.com

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ILOAD

RSENSE VBATTERY 2V to 25V

+

RS+

RS–

RGAIN

LOAD

RGAIN

2nd-Order Differential LPF fLP ≈ 50kHz

VDD +3.3V

+

– M1 PMOS

Microcontroller

TS1100 ROUT 10kΩ

OUT

ADC

0.047µF

GND Figure 1: A Typical Application for a High-precision Uni-directional Current-sense Amplifier (Touchstone Semiconductor’s TS1100).

error because of op-amp input bias current mismatch.

Bi-directional Current-Sense Amplifiers While uni-directional CSAs are primarily used in those applications where current is delivered to a load, there are many applications where it is necessary to measure current in both directions. Some applications where bi-directional current-sense monitoring/amplification are needed include: smart battery packs and chargers, portable computers, super capacitor charging/ discharging devices, and general-purpose currentshunt measurements.

Uni-directional CSAs were used prior to the advent of bi-directional CSAs; however, two uni-directional CSAs are necessary in order to measure current in both directions. Whereas the RS+/RS- input pair of CSA #1 is wired normally for measuring current to the load, the RS+/RS- input pair for CSA #2 would be wired antiphase with respect to CSA #1 for measuring current

28

back to the source. There are significant disadvantages to using this configuration: (a) the cost of two CSAs; (b) twice the printed-circuit-board (pcb) area is necessary because of the two CSAs; © two ADC inputs are consumed; and (d) additional microcontroller coding and machine cycles are required.

A straight-forward modification to the uni-directional CSA configuration yields a bi-directional CSA as shown in Figure 2 for Touchstone Semiconductor’s TS1101.

This implementation saves on additional computing resources, pcb area, and component costs.

The internal amplifier was reconfigured for fully differential input/output operation and a second lowthreshold p-channel FET (M2) was added as shown in Figure 2. The operation of this bi-directional CSA is identical to that of the uni-directional CSA previously discussed when VRS- > VRS+. When M1 is conducting current, the internal amplifier holds M2 OFF in the implementation shown in Figure 2. The amplifier holds

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M1 OFF when M2 is conducting current. The disabled FET does not contribute to the resultant output voltage in either case.

For both types of uni-directional or bi-directional CSAs, gain error accuracy is a measure of how well-controlled is the ratio of ROUT to RGAIN, especially over temperature. Gain error accuracy can be <0.5% using novel circuit techniques in monolithic implementations.

In a discrete CSA circuit, it would be quite difficult to achieve this level of performance over temperature with standard 1% tolerance and 100ppm/°C temperature coefficient resistors. Many CSAs offer different gain options tailored to specific application requirements while some CSAs are only available with fixed-gain options.

To achieve their very-low VOS performance over temperature, over wide VSENSE voltages, and over wide power supply voltages, higher-performance CSA incorporate chopper stabilization into the input stage – a commonly-used technique to reduce significantly amplifier VOS. Load currents can be resolved to 12-bit resolution or better for full-scale VSENSE voltages equal to and larger than 123mV in reducing the

VBATTERY 2V to 25V

CSAs’ VOSs to 30µV (typically) or less. Load current measurements are 2 times more accurate using CSAs that have implemented chopper-stabilized input stages when compared to similar CSAs that exhibit VOSs > 100µV or more.

The CSA’s SIGN Output Comparator The bi-directional CSA incorporated one additional feature as was shown in Figure 2 – an analog comparator the inputs of which monitor the internal amplifier’s differential output voltage. The SIGN comparator output indicates the load current’s direction while the voltage at its OUT terminal indicates the magnitude of the load current. The SIGN output is a logic high when M1 is conducting current (VRS+ > VRS-). Alternatively, the SIGN output is a logic low when M2 is conducting current (VRS+ < VRS-).

Note that the SIGN comparator exhibits no “dead zone” at ILOAD switchover, unlike other bi-directional CSAs where hysteresis was purposely introduced to prevent comparator output voltage chatter. The load current transition band is less than ±0.2mA with respect to a 50-mΩ external sense resistor. Other types of CSAs

ILOAD

RSENSE

To AC Wall Cube OR Charger

+

RS+

RS–

RGAINA

LOAD

RGAINB

2nd-Order Differential LPF fLP ≈ 50kHz

–

AMP +

M1 PMOS

VDD +3.3V

+

TS1100 1

TECH ARTICLE

–

M2 PMOS

OUT VDD +

ROUT 10kΩ

COMP

SIGN

Microcontroller

ADC Input Digital Input

–

0.1µF

0.047µF

GND Figure 2: A Typical Application for a Bi-directional High-precision Current Sense Amplifier (Touchstone Semiconductor’s TS1101).

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that also utilize an analog OUT/comparator SIGN arrangement exhibit a SIGN transition band that can range up to 2mV (or 40mA referred to a 50mΩ sense resistor). Low-transition band, bi-directional CSAs can be 200 times more sensitive on this attribute alone.

CSAs are specified to operate over wide or extended industrial temperature ranges, can operate from 2V to 25V (and higher) power supplies, and mate their electrical performance with pcb-space saving packages (such as SOT23-5 and SOT23-6).

Internal Noise Filters

About the Author

It’s always been good engineering practice to add external low-pass filters (LPFs) in series with the CSA’s inputs to counter the effects of externally-injected differential and common-mode noise prevalent in any load current measurement scheme. Resistors used in the external LPFs in the design of discrete CSAs were incorporated into the circuit’s overall design so errors because of any input-bias current-generated voltage and gain errors were compensated.

Adolfo Garcia has over 30 years of experience in the analog IC business. He has held design, applications, marketing, and product line/business unit management positions of increasing responsibility at Analog Devices, Linear Technology, Micrel, Advanced Analogic Technologies, and Leadis Technology. His technical and market knowledge spans a broad spectrum of analog products and applications, including amplifiers, data converters, and power management functions. This expertise has led to the definition and market launch of over 65 high-performance analog IC products. He holds three US Patents and is an accomplished author with over 60 publications to his credit.

Utilizing external LPFs in series with the CSA’s inputs only introduces additional offset voltage and gain errors with the advent of monolithic CSAs. Higherperformance uni-directional and bi-directional CSAs incorporate internal LPFs to further save system cost and improve overall system performance, thereby minimizing/eliminating the need for external LPFs and to maintain low offset voltage and gain errors.

Additional Applications Tips All parasitic pcb track resistances to the sense resistor should be minimized for optimal VSENSE accuracy. Strongly recommended are Kelvin-sense pcb connections between RSENSE and the CSAs’ RS+ and RS- terminals. The pcb layout should be balanced and symmetrical to minimize wiring-induced errors.

Also, the pcb layout for RSENSE should include good thermal management techniques for optimal RSENSE power dissipation.

To form an LPF with the CSAs’ ROUT, a 22nF to 100nF good-quality ceramic capacitor should be connected from the OUT terminal to GND. The use of a capacitor at this terminal minimizes voltage droop (holding VOUT constant during the sample interval). Using a capacitor on the OUT terminal will also reduce the CSAs’ smallsignal bandwidth as well as band-limiting amplifier noise.

A new state of the art in CSA technology has been redefined. These new CSAs can resolve charging or discharging currents with 12-bit or better resolution, exhibit very low VOS and gain match errors, are extremely easy to use, are self-powered, and consume very little supply current. These higher-performance

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