Three phase rectifier

View with images and charts Reduction of Harmonics of Input Side Currents of a Three Phase Rectifier Combining Active and Passive Filters.

Chapter 1 1.1 Introduction Since most loads in modern electrical distribution systems are inductive, there is an ongoing interest in improving power factor [1]. The low power factor of inductive loads robs a system of capacity and can adversely affect voltage level. To improve input power factor of power converter, normally a power factor correction (PFC) circuit is designed and placed in front end of the converter, which in turn interfaced with the load. This power factor correction circuit may be an independent unit, by which the power supply followed, or an inseparable part of circuit incorporated into the power supply of the load. The line is a voltage source and will not be distorted if the line current is a sinusoidal one. Hence the basic idea of doing PFC is straightforward, by certain means, to force the line current to follow the waveform of the line voltage. However, there exists an unbalance of instantaneous power between the input powers of the PFC circuit, which is an alternative quantity with two times the line frequency, and its dc output power. Therefore, the operation principle of a PFC circuit is to process the input power in certain way that it stores the excessive input energy when the input power is larger than the dc output power, and releases the stored energy when the input power is less than the dc output power. To accomplish the above process, at least one energy storage element must be included in the PFC circuit. Various active and passive techniques are being carried out to meet the stander. To improve the power factor and to comply with various national and international line-harmonics regulations in lineoperated electronic equipment, an active or passive power-factor-correction/lineharmonics-reduction circuit must be added to the capacitive filtered bridge or voltagedoublers front-end rectifier. Although it is straightforward to obtain negligible distortion (i.e. less than 5%) and high power factor (more than 99%) with active circuits operating at high frequency (above 20 kHz), the additional circuitry can significantly reduce the reliability and increase the complexity, EMI, and Cost of the equipment. Passive solutions (circuits without controlled switches) offer an attractive trade-off between cost and performance: they are simple, reliable, robust, generate no EMI, and still provide compliance with the norms 1.2

Definiton

Power factor correction is simply defined as the ratio of real power to apparent power, or:

PF = Real Power / Apparent Power

Where, the real power is the average, over a cycle, of the instantaneous product of current and voltage, and the apparent power is the product of the rms value of current times the rms value of voltage. If both current and voltage are sinusoidal and in phase, the power factor is 1.0. If both are sinusoidal but not in phase, the power factor is the cosine of the phase angle.

1.3 Background

The new European Norms EN 60555 and the international standard IEC555 will impose a limit on the harmonic content of the input current of mains supplied equipment. In practice this will require the addition of a Power Factor Corrector (PFC) at the input of many types of mains operated electronic equipment, for example electronic lamp ballasts, TV power supplies and motor drives. A correctly designed PFC draws a sinusoidal input current from the mains supply, in phase with the mains voltage, and meets the EN60555 norm. It may also provide additional functions, such as automatic mains voltage selection and a regulation of the voltage supplied to the attached equipment [2].

Passive techniques, which introduce a filtering stage consisting of inductor and/or capacitor, to reduce low frequency harmonics, are attractive for their simplicity and reliability. But size and weight are their major drawback. Active techniques on the other hand, use a high switching frequency converter that shapes the input currents almost sinusoidal with small harmonic contents.

That is why the use of active filter is getting increasing interest.

In most PFC circuits, an input inductor has been connected to the bridge rectifier. Because of the current continuity nature of inductor, we usually call such connection as “current driven”.

The input inductor can operate in either continuous conduction mode (CCM) or discontinuous conduction mode (DCM). In DCM, the input inductor in no longer a state variable since its state in a given switching cycle is independent on the value in the previous switching cycle. The peak of the inductor current is sampling the line voltage automatically.

This property of DCM input can be called “self-power factor correction” because no control loop is required from its input side.

Among the topologies shown in figure 2, the boost configuration operating in a continuous current mode (ie the value of the inductor at the input is calculated such that it conducts continuously throughout the switching cycle) applies the smallest amount of high frequency current to the input capacitor Ci . It is the only topology which allows the noise across the input capacitor to be reduced, which is the major factor defining the size and cost of the filter.

Additionally, the boost inductor stores only a part of the transferred energy (because the mains still supplies energy during the inductor demagnetization) and so the inductor required is smaller in comparison with the other topologies. The boost topology thus leads to the cheapest PFC solution, but does not provide either in-rush current or short circuit protection.

The buck/boost topology can also be used; its advantages are that it can provide output isolation and adjustable output voltage. This paper will take the cost as the most important consideration, and so will concentrate on the boost circuit topology.

Figure 1.1 Active PFC topologies 1.4 Objective of the work The objective of this thesis is to make the input current of a three-phase rectifier circuit to be nearly sinusoidal and at the same time in phase with the supply potentials. This eliminates the harmonic contents in the input current and improves the power factor of the circuit. In doing so, the efficiency of the module is also improved. As a whole, the size and ratings of the equipments has been minimized. The aim of the study is to investigate an integrated power quality improvement by simultaneous active and passive filtering technique.

In this work, the appropriate switching frequency and solid-state switch has been selected to shape the input current. But this leads to introduce very high frequency harmonics in the wave. To eliminate these high frequency components in the input, a simple filter with very small rating inductor and capacitor is used to bypass these high frequency components.

However, even after use of input filter low frequency components still exists in the input current.

That is why another series LX filter has been set up at the input side. The component values of this input filter are modeled in such a way that they would resonate at the supply frequency so that power factor of the circuit remains near unity. All of this modeling has been done by using the simulation and the mathematical expression for the acceptable range of values of the components are found out.

1.5

Thesis Layout

Organization of this thesis includes seven chapters. Chapter one gives a general introduction followed by background & objective of the work.

Chapter two reviews basic concepts of power factor correction & classification of active, passive approach

In chapter three, different rectifier circuit with passive & active filters are analyzed to find out a perfect solution of our research. Simulation of each circuit is also provided.

Chapter four includes Comparison of Basic Converter Topologies For Power Factor Correction.

Also include Cuk converter, Sepic converter and Zeta converter Chapter five includes the proposed scheme that has to be implemented. It has four stages.

First stage is a resonating low frequency filter, second stage has high frequency filter designed by high frequency transfer function, third stage is an IGBT switch that is driven by a separate switching module & the last & forth stage is load with capacitive filter for reducing the output voltage ripples.

Conclusive discussions & remarks are drawn in chapter six . Some suggestions leading to future scope of work are also presented here.

An appendix is included at the end. It describes Power Factor Controller.

Chapter 2 BASIC CONCEPT OF POWER FACTOR CORRECTION 2.1

Power factor

Power factor (pf) is defined as the ratio of the real power (P) to apparent power (S), or the cosine (for pure sine wave for both current and voltage) that represents the phase angle between the current and voltage waveforms (see Figure 1). The power factor can vary between 0 and 1, and can be either inductive (lagging, pointing up) or capacitive (leading, pointing down). In order to reduce an inductive lag, capacitors are added until pf equals 1.

When the current and voltage waveforms are in phase, the power factor is 1 (cos (0째) = 1).

The whole purpose of making the power factor equal to one is to make the circuit look purely resistive (apparent power equal to real power). Real power (watts) produces real work; this is the energy transfer component (example electricity-to-motor rpm). Reactive power is the power required to produce the magnetic fields (lost power) to enable the real work to be done, where apparent power is considered the total power that the power company supplies, as shown in Figure 1. This total power is the power supplied through the power mains to produce the required amount of real power.

Figure 2.1 : Power Factor Triangle (Lagging)

The previously-stated definition of power factor related to phase angle is valid when considering ideal sinusoidal waveforms for both current and voltage; however, most power supplies draw a non-sinusoidal current. When the current is not sinusoidal and the voltage is sinusoidal, the power factor consists of two factors: 1) the displacement factor related to phase angle and 2) the distortion factor related to wave shape.

5 Equation 1 represents the relationship of the displacement and distortion factor as it pertains to power factor. ………….(1) Irms(1) is the current’s fundamental component and Irms is the current’s RMS value. Therefore, the purpose of the power factor correction circuit is to minimize the input current distortion and make the current in phase with the voltage.

When the power factor is not equal to 1, the current waveform does not follow the voltage waveform. This results not only in power losses, but may also cause harmonics that travel down the neutral line and disrupt other devices connected to the line. The closer the power factor is to 1, the closer the current harmonics will be to zero since all the power is contained in the fundamental frequency.

2.2 Harmonics Harmonics are currents or voltages with frequencies that are integer multiples of the fundamental power frequency. In the U.S., the fundamental frequency is 60Hz. Harmonics are created by non-linear loads that draw current in abrupt pulses rather than in a smooth sinusoidal manner.The effects of harmonics can be overheating of transformers, cables, motors, generators and capacitors connected to the same power supply with the devices generating the harmonics. Electronic displays and lighting may flicker, circuit breakers can trip, computers may fail and metering can give false readings.

2.3 Need of Power Factor Correction The current flow through the circuit is increased by the reactive component. Normally, loads are represented by a series combination of a resistance and a purely imaginary reactance. For this explanation, it is easier to contemplate it as an equivalent parallel combination. The diagram below illustrates a partially reactive load being fed from a real system with some finite resistance in the conductors, etc.

6

The current through the reactive component (Ireactive) dissipates no power, and neither does it register on the watt hour meter. However, the reactive current does dissipate power when flowing through other resistive components in the system, like the wires, the switches, and the lossy part of a transformer (Rline). Switches have to interrupt the total current, not just the active component. Wires have to be big enough to carry the entire current, etc. Correcting the power factor reduces the amount of over sizing necessary.

In elementary courses in electricity, this is sometimes taught as the definition of power factor, but it applies only in the special case, where both the current and voltage are pure sine waves. This occurs when the load is composed of resistive, capacitive and inductive elements and all are linear (invariant with current and voltage). Switched−mode power supplies present non−linear impedance to the mains, as a result of the input circuitry. The input circuit usually consists of a half−wave or full−wave rectifier followed by a storage capacitor. The capacitor maintains a voltage of approximately the peak voltage of the input sine wave until the next peak comes along to recharge it. In this case, current is drawn from the input only at the peaks of the input waveform, and this pulse of current must contain enough energy to sustain the load until the next peak. It does this by dumping a large charge into the capacitor during a short time, after which the capacitor slowly discharges the energy into the load until the cycle repeats. It is not unusual for the current pulse to be 10% to 20% of the cycle duration, meaning that the current during the pulse must be 5 to 10 times the average current in magnitude. Figure 2.2 and 2.3 illustrate this situation.

Figure 2.2 : Input Characteristic of a Typical Switched-Mode Power Supply without PFC

Figure 2.3 : Harmonic Content of the Current Waveform without PFC

Note that the current and voltage can be perfectly in phase, in spite of the severe distortion of the current waveform. Applying the “cosine of the phase angle” definition would lead to the erroneous conclusion that this power supply has a power factor of 1.0.

The input of a power supply with perfect power factor correction. It has a current waveform that mimics the voltage waveform, both in shape and in phase. Note that it’s input current harmonics are nearly zero.

Figure 2.4 : Input Characteristic of a Power Supply with Near-Perfect PFC 2.4 Process of power factor Correction Given the reactive load component (Xload), you can calculate the capacitance that would be put in parallel to exactly match it using the equation: Xc = 1/ (omega C) = 1/(2 *pi * f * for 60 Hz: Xc = 1/( 2*pi * 60* C) =1/ (377*C) or, rearranging: C = 1/(377*Xc)

C)

Power factor correction capacitors are often rated in kVar, instead of uF, because that is how the power company works. Say a factory has several thousand horsepower worth of motors at .85 power factor. They might have a reactive component of several hundred kVar. At a distribution voltage of 14,400 volts, this would require a capacitor with an impedance of a bit more than 1000 ohms, or about 2.5 microfarads, a reasonable sized and priced package.

However, if you were crazy enough to try to compensate this at 230 volts, you would need about .01 Farads (i.e. 10,000 uF), a sizeable package.

For very large systems, even capacitors get unwieldy. One approach is to use large over excited synchronous motors which look like capacitors, electrically. Another approach is clever systems of thyristors and inductors which simulate the capactive reactance by drawing "displacement current".

2.5 Power Factor Correction vs. Harmonic Reduction It is clear from the previous illustrations that high power factor and low harmonics go hand−in−hand. There is not a direct correlation however, the following equations link total harmonic distortion to power factor.

Where Kd is the distortion factor and is equal to –

Therefore, when the fundamental component of the input current is in phase with the input voltage, K = 1 and:

As illustrated, a perfectly sinusoidal current could have a poor power factor, simply by having its phase not in line with the voltage.

A 10% Total Harmonic Distortion (THD) corresponds then to a Power Factor approximately equal to 0.995. It is clear that specifying limits for each of the harmonics will do a better job of controlling the “pollution” of the input current, both from the standpoint of minimizing the current and reducing interference with other equipment. So, while the process of shaping this input current is commonly called “power factor correction,” the measure of its success in the case of the international regulations is the harmonic content.

2.6 Loads that draw non-sinusoidal current Classic reactive loads, like transformers, lighting ballasts, and AC motors still have a sinusoidal current flow. The phase of the current is just shifted from that of the supply voltage. However, there are some loads which draw distinctly non-sinusoidal currents. The most recently notorious is the switching power supply in a PC. These power supplies start with a bridge rectifier feeding a capacitor, and so, particularly at part load, draw their current in little peaks, when the instantaneous line voltage is above the capacitor voltage, forward biasing the rectifier.

Another notorious non-sinusoidal current draw is the popular phase controlled light dimmer, which uses a TRIAC or SCR to reduce the RMS voltage to the load by turning on partway through the half cycle. Not only is the current waveform highly non-sinusoidal, but it is also out of phase with the voltage supply. Hence, these loads have a non-unity power factor, and draw reactive power.

However, to compensate these loads, you have to come up with a means to supply the reactive current at the appropriate times. A simple capacitor doesn't do this. A capacitor only compensates nice sinusoidal power factor lags, like those from linear (non-saturating) inductors.

2.7 Example of Power Factor Correction Let's take an example. A 3/4 HP electric motor has a power factor of .85. The nameplate current is 10 Amps at 115 Volts, or 1150 Volt Amps.

• • •

Apparent power = 1150 Volt Amps Active power (P) = .85 * 1150 = 977.5 Watts Reactive Power (Q) = sqrt(1150^2 - 977.5^2) = 605 VAR

So, we need about 600 var of power factor correction. I'm rounding to a couple digits, because, in reality, it's unlikely that the power factor is known to more accuracy, nor will any of the PFC components be that precise. (10% accuracy would be quite good for a capacitor).

Now, assume we want to put the capacitor in parallel with the motor: Calculating the required impedance from Q = E^2/X, where Q is the reactive power needed:

• •

600 = 115^2/X => X = 115^2/600 = 22 ohms (rounding to 2digits) C = 1 /( 2 * pi * f *X) = 1/ (377 * 22) = 120 uF (again, rounding to 2 digits)

which is a fairly large capacitor in a constant duty environment (i.e. motor run, as opposed to motor start, where the capacitor is only in the circuit for a short time). We can calculate the RMS current through the capacitor either by dividing the VARs by the line voltage (600/115) or by dividing line voltage by reactance (115/22); both come out at around 5 1/4 Amps, so we'd want a capacitor rated at somewhat more current (e.g. 7-10 A). The capacitor's series resistance should be pretty low, or it will dissipate a fair amount of energy. If the dissipation factor were 1%, we'd be dissipating about 6 Watts in the capacitor. One can also put the PFC capacitor in series with the load. In this case the capacitor would carry the entire load current of 10A, but, the required value is different.

For a series compensation, we'd determine the series equivalent of the load (we used a parallel model, above). For the series model, you use currents, instead of voltages: 600 VAR = I^2 * X => 600 = 10*10 * X => X = 6 ohms And converting an impedance to a capacitance: C=1/(377*6) = 440 uF. So, not only would the capacitor be larger, but it would need to carry the entire load current.

For this example, at least, parallel PFC seems to be a better approach. Only if the power factor were very poor, so the reactive impedance was quite large (and the corresponding capacitance low) would series compensation seem to be useful.

If the line voltage were higher, the correction impedance would be increased as the square of the line voltage. The capacitance would be reduced as the square of the line voltage. That is, if the same motor were run off 230 Volts, the capacitor would only need to be about 30 uF. And if we were to do power factor compensation at the distribution voltage of 4160 volts (for example), we would only need about .1 uF. This is why power factor correction is usually done in the distribution network at MV or HV, and not at the end voltage.

2.8 Classification of Power Factor Correction Approaches The general approaches to improve power factor can be widely classifieds as passive and active approaches[3]. The passive approaches use capacitive inductive filters to achieve PCF, while the active approaches use a switched mode power supply to shape the input current.

However, there are no rules demanding that the PFC be accomplished by active circuits (transistors, ICs, etc.). Any method of getting the harmonics below the regulatory limits is fair game. It turns out that one inductor, placed in the same location as the active circuit, can do the job.

Waveforms: 1. Input current with no PFC 2. Input current with passive PFC 3. Input current with active PFC 4. Input voltage

Figure 2.5 : Input Characteristics of PC Power Supplies with Different PFC Types (None, Passive, and Active) 2.8.1 Passive Approaches In the passive approaches, a full bridge rectifier with an LC filter is used to reduce the line current harmonic limits. Generally the LC filter can be placed in either the AC-side or the DC-side of the rectifier as shown in Figure 2.6. Placing the LC filter in the AC-side will result in more pure sinusoidal input current. Passive PFC can meet the regulation with high efficiency, superior reliability, low cost, and low EMI [4-5]. On the other hand, the filter capacitor voltage vanes with the line voltage, which has a detrimental effect on the performance and efficiency of the DC-DC converter. When considering a hold-up time for the power supply, the bulk capacitance has to be increased and becomes very bulky compared to what it would be without this varying voltage.

As a result, the passive approaches seem to be more attractive in low-power applications, up to 300Watts, and are more suitable for narrow line voltage range. Other drawbacks are the size and weight of the filter choke inductor.

However, the majority of power supplies manufactured in low-power and costsensitive applications have adopted the passive PFC approaches. .

Vac

Passive PFC Filter

Passive PFC Filter

DC-DC Converter

LOA D

Control Figure 2.6 : General Structures of the Passive PFC Approaches 2.8.2 Active Approaches In active PFC approaches, a switched mode converter is employed to overcome the limitations of the passive approaches. Assuming unity power factor, the line current should be sinusoidal and in phase with the line voltage. That will result in pulsating output power than contains - in addition to the real (average power) - an alternating component with double-line frequency.

Since the power demanded by most loads is constant, an energy storage element is needed.

Since the inductor-stored energy cannot match this excessive energy, another storage component is needed. This storage capacitor is normally located between the two stages and

should handle the double-line frequency ripple component, which make it bulky. This second, harmonic problem that presents itself on the output of the PCF stage cannot be internally solved. Usually, a compromise between PFC and output voltage ripple should be made, but most of the time this output voltage is not good enough to supply the load. As a result, another DC-DC converter is needed, or what is so called post regulator, to solve this problem and achieve tight output regulation. The result in the most powerful PFC configuration and is the active two-stage PFC shown in Figure 2.7.

2.8.2 (A) Two-Stage PFC converter This configuration implies the use of two converters to achieve both power factor correction and output regulation in addition to the rectification circuit and the input EMI filter. These converters are independent, which means each one has its own switches and control circuit.

The PFC converter performs the input current shaping using one of the popular converter topologies (buck, boost, buck-boost, flyback, SEPIC, Cuk, ZETA) in addition to one of the PFC controlling techniques. The boost converter is widely used due to its advantages, which include good power factor, grounded switch, input inductor and simplicity. However, this PFC converter normally has a low bandwidth control, which implies a loosely regulated output voltage across the storage capacitor. In universal line voltage applications, the DC bus voltage may vary between 380-400V. Because of the relative high voltage on the storage capacitor, the value of the capacitance can be optimized to provide the necessary hold up time. The DC-DC converter is connected to the storage capacitor to provide the necessary tight output voltage regulation with the appropriate gain and, most of the time, provides isolation.

Vac

PFC Converter

DC-DC CS

Control

Converter

LOAD

Control

Figure 2.7 : System Configuration of Two-Stage PFC Power Supply (B) Single-Stage PFC converter The single-stage PFC configuration came about to reduce the cost and complexity of the twostage structure, and it can be viewed more as a modification on the two stage PFC rather than a class by itself. As can be seen from Figure 2.8, the PFC and the DC-DC cell share the control circuit and can also share the switches in this configuration. The energy storage capacitor between the two stages serves as a buffer for frequency isolation and to provide the converter with the necessary hold up time. However, in single stage configuration, the voltage across the storage capacitor is not regulated, -because the controller is used to

regulate the output voltage. As a result, this voltage can vary greatly, usually between 130500V in universal line application. This will have a negative impact on the design and cost of the PFC converter.

PFC Cell

Vac

DC-DC Cell LOAD

CS

Control Figure 2.8 : System Configuration of Single-Stage PFC Power Supply 2.9 Approach Comparisons Generally, the passive approach should be considered in low power applications, especially when designing to meet the minimum regulation requirements with a narrow line voltage range. At low power levels (<300W), the active single-stage approach offers a great advantage over the passive approaches due to its simple structure, low cost, minimum weight and better PFC performance.

Unity power factor and tight output regulation for any power range can be achieved through the two-stage active PFC. This structure can guarantee compliance with any regulation and is compatible with universal line voltage applications. Some negative factors of the two-stage scheme include cost, size, and sometimes its lower efficiency.

Table-1 Provides a general relative performance comparison for the passive and active single- and two stage approaches [6] .

Performance Review THD Power Factor Efficiency Size Bulk-Cap Voltage Control Component Count Power Range

Passive Scheme High Low High Large Variation Simple Least <300 W

Chapter 3 IMPROVEMENT OF THD RECTIFIER

Active Two- Stage Low High Medium Medium Constant Complex Medium Any

Active Single- Stage Medium Medium Low Medium Small Variation Medium Medium-Low <300 w

3.1 Analysis of 3-Phase rectifier circuit Three phase diode rectifier circuits are extensively used in many high power low cost applications leading the degradation in the power quality due to the current distortion. A simple three phase rectifier circuit and its input current and harmonics are shown in the figure 3.1, 3.2, 3.3 respectively.

Figure 3.1: A simple diode bridge rectifier without capacitor Here input voltage V1, V2, V3 peak value 300v with phase difference 120 deg. Frequency 50Hz, R1 100â&#x201E;Ś.

Figure 3.2a: Input voltages of the three phases of the rectifier for Phase A, B, C

Figure 3.2b: Input currents of the three phases of the rectifier for Phase A, B, C

Figure 3.3: Harmonic content of the three phases A, B, C Table 3.1: Harmonic content of Current in the phase A without capacitor Harmonics

Values (mA)

Harmonics

Values (mA)

I-1(50Hz)

5500

I-11(550Hz)

I-2(100Hz)

45

I-12(600Hz)

38

I-3(150Hz)

20

I-13(650Hz)

75

I-4(200Hz)

44

I-14(700Hz)

45

I-5(250Hz)

1100

I-15(750Hz)

12

I-6(300Hz)

55

I-16(800Hz)

38

I-7(350Hz)

500

I-17(850Hz)

100

I-8(400Hz)

48

I-18(900Hz)

18

300

I-9(450Hz)

6

I-19(950Hz)

52.5

I-10(500Hz)

42

I-20(1000Hz)

36

√∑ (Mh ) 2 THD% =————

* 100% M1

Where Mh is the magnitude of either voltage or current harmonic component and M1is the magnitude of either the fundamental voltage or current.

Putting the values in the equation we have got the THD values for a simple rectifier is 22.9%.

18

Figure 3.4:Output voltage of this rectifier without capacitor

Figure 3.5: Phase relation of input current and output voltage without capacitor

3.2 A three diode bridge rectifier with capacitor

Figure 3.6: A simple diode bridge rectifier with capacitor

Figure 3.7: Input currents of the three phases of the rectifier with capacitor Table 3.2: Harmonic content of Current in the phase A with capacitor Harmonics

Values (mA)

Harmonics

Values (mA)

I-1(50Hz)

5500

I-11(550Hz)

1175

I-2(100Hz)

50

I-12(600Hz)

43

I-3(150Hz)

33.6

I-13(650Hz)

1050

I-4(200Hz)

32

I-14(700Hz)

16.5

I-5(250Hz)

3450

I-15(750Hz)

75

I-6(300Hz)

16

I-16(800Hz)

47

I-7(350Hz)

2150

I-17(850Hz)

750

I-8(400Hz)

35

I-18(900Hz)

53

I-9(450Hz)

77.5

I-19(950Hz)

590

I-10(500Hz)

60

I-20(1000Hz)

21

The THD value obtained from this rectifier is 83.23%. This is an extremely large value.

Because of the insertion of the capacitor at the o/p to make the o/p volt. Ripple free the i/p current becomes too much distorted and harmonic content has increases a lot

Figure 3.9: Output voltage of the rectifier with capacitor

Figure 3.10: Phase relation of i/p current and o/p Volt. with capacitor

3.3 Harmonic reduction with passive filter By observing the input current wave shape of these filters we can say about the harmonic contents of them. There is no even harmonics as the waveforms are symmetrical about the Xaxis. Another noteworthy fact is balanced three phase rectifier type loads do not produce a third harmonic component. Nor do they produce any triplet harmonic component. Again the 11th harmonic & higher is a point where the magnitude diminishes to a very low level. Thus 5th & 7th orders are the problem child harmonics for AC drives.

Table 3.3: Harmonic spectrums Analysis Harmonic fundamental 5th 7th 11th 13th 17th 19th

Value/unit 1 0.2 0.14 0.09 0.07 0.06 0.05

Frequency(Hz) 50 300 350 550 650 850 950

Passive filters may be used for reducing the harmonics content of the output currents. But they do not allow the regulation of the output voltage & also decreases the output voltage levels in comparison with the unfiltered rectifiers.

Taking the harmonic limits as a quality index, the resulting inductors are typically larger than the ones used in high quality using circuits[7-8].

3.4 Effect of input inductance

Figure 3.11: Three-phase rectifier with input inductance

Figure 3.12: Input current with 1mH inductance

Figure 3.13: Output voltage with 1mH inductor

Figure 3.14: Input current with L1=L2=L3=100mH

Figure 3.15: Output voltage with L1=L2=L3=100mH

Figure 3.16: Input current with L1=L2=L3=1H

Figure 3.17: Output voltage with L1=L2=L3=1H As we can see with increasing the inductive value of the i/p inductor the current wave shape improves but the o/p voltage decreases.

3.5 Rectifier Analysis with passive filter

Figure 3.18: Rectifier with i/p passive filter

Figure 3.19: Input current & Output voltage L1=L2=L3=100mH, C1=C2=C3=100uF

Figure 3.20: Input current & Output voltage L1=L2=L3=1H, C1=C2=C3=10uF

Figure 3.21: Input current & Output voltage L1=L2=L3=10mH, C1=C2=C3=100uF

Figure 3.22: Input current & Output voltage L1=L2=L3=10mH, C1=C2=C3=10uF

Figure 3.23: Input current & Output voltage L1=L2=L3=10H, C1=C2=C3=100uF

Figure 3.24: Input current & Output voltage L1=L2=L3=100mH, C1=C2=C3=10uF From the wave shapes shown above, we see that the value of the inductor increases the input wave shapes is improving a lot but output voltage level decreases & decreasing the value of inductor given more output voltage but input current distortion increases.

The best model for the passive filter has obtained when L=100mH & C=100uF. Actually the filter can be modeled by calculating the resonating values as XL=XC.

Calculation is done by considering the fundamental component as 50Hz. The product of LC should be 1*10-5 . So in a passive input filter the component values are very large & regulation of the output voltage is not possible.

3.6 Introduction of switching in Boost Rectifier Active wave shaping means to introduce switching in the rectifier circuit. In a boost rectifier if switching is introduced without having any input filter the input current start to conduct in Discontinuous Conduction Mode (DCM) & the wave shape follows the input voltage but have high frequency switching components. The output cannot reach to a desired level as expected by the boost converter. So, though the input current wave shape has improved a lot, the efficiency of this module is not good at all. In Figure 3.26 the input current & output voltage wave shapes are given that we got from simulation.

Figure 3.25: A single switch boost rectifier

Figure 3.26: PWM circuit.

31

Figure 3.27: The input voltage (Vpulse) of the OpAmp at pin#3 A 4kHz saw tooth wave varying from 15V to 0V.

Figure 3.28: Input current & Output voltage with switching With the introduction of EMI filter at the input can make better the performance. The circuit model for switching with input filters where L1, L2, & L3 are of 500uH. Here we can see the output voltage has got to an acceptable high value as expected by boost rectifier.

3.7 Boost Rectifier with EMI filter and Switching

Figure 3.29: A three phase boost rectifier with single switch and i/p filter

Figure 3.30: Duty Cycle of PWM Duty Cycle 0.17 V1 =V2=V3= 300volt.(peak)

Figure 3.31: Input current for boost rectifier with i/p filter and switching

Figure 3.32: Output voltage for boost rectifier with i/p filter and switching

Figure 3.33: Input current Harmonics content of the phases A for boost rectifier with i/p filter & switching Table 3.4: Harmonic content of Current in the phase A with switching & EMI filter Harmonics

Values (mA)

Harmonics

Values (mA)

I-1(50Hz)

820000

I-11(550Hz)

6250

I-2(100Hz)

I-12(600Hz)

I-3(150Hz)

I-13(650Hz)

I-4(200Hz)

I-14(700Hz)

I-5(250Hz)

50000

I-6(300Hz) I-7(350Hz)

1562.5

I-16(800Hz) 25000

I-8(400Hz) I-9(450Hz)

I-15(750Hz)

3125

I-17(850Hz)

7811

I-18(900Hz) 12500

I-10(500Hz)

I-19(950Hz)

390

I-20(1000Hz)

A THD of a three-phase rectifier with capacitor can be reduced from 83.23% to 17.61% with the introduction of active filter or switching in the rectifier. Even at this value of the THD is not acceptable as recommended by different regulations & standards. Some application requires the THD to be less than 10% & even some other application have set the limit to be less than 5%. That is why research is still going on to improving the THD level. In our work, we tried to reduce the value to be less than 5%.

Duty Cycle 0.435 V1 =V2=V3= 17volt.(peak)

Figure 3.34: Input current for boost rectifier with i/p filter and switching

Figure 3.35: Output voltage for boost rectifier with i/p filter and switching

Figure 3.36: Input current Harmonics content of the phases A for boost rectifier with i/p filter & switching Table 3.5: Harmonic content of Current in the phase A with switching & EMI filter Harmonics

Values (mA)

Harmonics

Values (mA)

I-1(50Hz)

3250

I-11(550Hz)

50

I-2(100Hz)

I-12(600Hz)

I-3(150Hz)

I-13(650Hz)

I-4(200Hz)

I-14(700Hz)

I-5(250Hz)

200

I-15(750Hz)

I-6(300Hz)

I-16(800Hz)

I-7(350Hz)

I-17(850Hz)

25 12.2 6.25

I-8(400Hz) I-9(450Hz) I-10(500Hz)

I-18(900Hz) 100

I-19(950Hz)

3.125

I-20(1000Hz)

Chapter 4 Comparison of Basic Convertor Topologies For Power Factor Correction 4.1 Input Voltage-Current Characteristics of Basic Converter Topologies In order to examine the self-PFC capabilities of the basic converters, we first investigate their input characteristics. Because the input currents of these converters are discrete when they are operating in DCM, only averaged input currents are considered. Since switching frequency is much higher than the line frequency, let's assume the line voltage is constant in a switching cycle. In steady state operation, the output voltage is nearly constant and the variation in duty ratio is slight. Therefore, constant duty ratio is considered in deriving the input characteristics. During the analysis, the following nomenclatures are used : vl(t)=Vim sinωit — line voltage; il(t) — line current; vl(t) — rectified line voltage; il(t) — rectified line current; il,avg(t) — average value of rectified line current, (assume il(t) = il,avg(t)); V0 — output dc voltage; Ωl— line angular frequency; Tl— line period; Ts — switching period; D — duty ratio; Dl — input inductor discharging time ratio.

A. Buck converter The basic buck converter topology and its input current waveform when operating in DCM are shown in Figure 4.1(a) and 4.1(b), respectively. It can be shown that the average input current in one switching cycle is given by,

38

Figure 4.1(c) shows that the input voltage-input current V-I characteristics is a straight line. It should be note that this straight line does not go through the original. When the rectified line voltage vl(t) is less than the output voltage Vo negative input current would occur. This is not allowed because the bridge rectifier will block the negative current. As a result, the input current is zero near the zero cross point of the line voltage, as shown in Figure 4.1(c).

Actually, the input current is distorted simply because the buck converter can work only under the condition that the input voltage is larger than the output voltage. Therefore the basic buck is not a good candidate for DCM input power factor correction.

Figure 4.1: Input V-I Characteristic of basic buck Converter operating in DCM B. Boost converter The basic boost converter and its input current waveform are shown in Figure 4.2(a) and 4.2(b), respectively. The input V-I characteristic can be found as follows:

By plotting Eq. (2), we obtain the input V-I characteristic curve as given in Figure 4.2(c). As we can see that as long as the output voltage is larger than the peak value of the line voltage in certain extent (depending on Dl), the relationship between vl(t) and il,avg(t) is nearly linear.

When the boost converter connected to the line, it will draw almost sinusoidal average input current from the line, when the output voltage is higher than its input shown as in Figure 4.2(c).

Figure 4.2: Input V-I Characteristic of basic boost converter operating in DCM As one might notice from Eq. (2) that the main reason to cause the non-linearity is the existence of Dl. Ideally, if Dl = 0, the input V-I characteristic will be a linear one. In practice, to reduce the discharge period Dl, by properly configuring the circuit topology, a higher voltage, instead of V0,can be created to be applied to the inductor during Dl to discharge the inductor[11].

Because of the above reasons, boost converter is comparably superior to most of the other converters when applied to do PFC[11-13]. However, it should be noted that boost converter can operate properly only voltage. When low voltage output is needed, a step-down dc-dc converter must be cascaded.

C. Buck-boost converter Figure 4.3(a) shows a basic buck-boost converter. The averaged input current of this converter can be found according to its input current waveform, shown in Figure 4.3(b).

Figure 4.3 Input V-I Characteristic and input waveforms.

Equation (3) gives a perfect linear relationship between il.avg(t) and vl(i), which proves that a buck-boost has excellent self-PFC property. This is because the input current of buck-boost converter does not related to the discharging period Dl. Its input V-I characteristics and input voltage and current waveforms are shown in Figure 4.3(c). Furthermore, because the output voltage of buck-boost converter can be either larger or smaller than the input voltage, it demonstrates strong availability for DCM input technique to achieve power factor correction.

So, theoretically buck-boost converter is a perfect candidate. Unfortunately, this topology has two limitations: 1) the polarity of its output voltage is reversed, i.e., the input voltage and the output voltage don't have a common ground; and 2) it needs floating drive for the power switch. The first limitation circumscribes this circuit into a very narrow scope of applications.

As a result, it is not widely used.

D. Flyback converter

Figure 4.4 Input V-I Characteristic of basic flyback converter operating in DCM 42 Flyback converter is an isolation converter. Its topology is shown in Figure 4.4(a). Figure 4.4(b) shows its input current waveform. The input voltage-input current relationship is similar to that of buck-boost converter:

Where, Lm is the magnetizing inductance of the output transformer. Therefore, it has the same input V-I characteristic, and hence the same input voltage and input current waveforms as those the buck-boost converter has, shown in Fig. 4(c). Comparing with buck-boost converter, flyback converter has all the advantages of the buck-boost converter without any limitation. What's more, input-output isolation can be provide by flyback converter. These advantages make flyback converter most preferable in power factor correction with DCM input technique [14-16].

E. Forward converter

Figure 4.5 Forward converter and its input current waveform The circuit shown in Figure 4.5 is a forward converter. In order to avoid transformer saturation, it is well know that forward converter needs the 3 rd winding to emagnetize (reset) the transformer. When a forward converter is connected to the rectified line voltage, the demagnetizing current through the 3 rd winding is blocked by the rectifier diodes. Therefore, forward converter is not available for PFC purpose.

F. Cuk converter, Sepic converter and Zeta converter It can be shown that Cuk converter, Sepic converter and Zeta converter given in Figure 4.6(a), (b) and (c), respectively, have the same input V-I characteristic. Each of these converter topologies has two inductors, with one located at its input and the other one at its output. Let's consider the case when the input inductor operates in DCM while the output inductor operates in CCM. To investigate the input characteristic of these converters, let's take the Cuk converter as an example. One should note that the results from the Cuk converter are also suitable for Sepic converter and Zeta converter.

Figure 4.6 : Basic topologies of Cuk, Sepic and Zeta converters The waveforms of the Cuk converter, shown in Figure 4.6(a), for input inductor current (=input current), output inductor current and the current through the capacitor C are depicted in Figure 4.7(a). Assume that the output current is I 0. Then the average output inductor current is I0. In steady state, employing charge equilibrium principle, we obtain

44 Then the averaged input current can be found as

Figure 4.7 : Input V-I Characteristic of basic Cuk converter operating in DCM input From Eq.(6), the input V-I characteristic of Cuk converter is plotted in Fig. 7(b). According to this input V-I chart, the input current waveform corresponding to a sinusoidal input voltage is sketched. As we can see that the input current waveform is a distorted one. Therefore, Cuk converter does not have a good self-PFC property. This conclusion can be also extended to the Sepic and Zeta converters.

4.2 Conclusions According to the above discussion, we may conclude that the basic boost converter, flyback converter and buck-boost converter have excellent self-PFC capability naturally. Among them, boost converter and flyback converter are especially suitable for DCM PFC usage.

Hence, these two converters are most preferable by the designers for power factor topologies purpose. Other converters may be used only if their input V-I characteristics have been modified (linearized), or when they operate in continuous conduction mode. The characteristics of the above eight basic converter topologies are summarized as in Table I. Table 4.1 Comparison of various kinds of Topology

Chapter 5 ANALYSIS OF THE PROPOSED MODEL 5.1 Introduction The Buck and Boost inductors are most commonly used in “switchmode” or feedforward type of converter configurations. A forward type converter is said to be operating in “continuous mode” as long as the current through the inductor remains above zero for the entire switching cycle. If, at minimum output current, the current through the inductor drops to and remains at zero during some portion of the cycle, the converter is being operated in the “discontinuous mode.” A real world example of this application would be in a battery charger for your laptop.

5.2 Buck Inductor design So, when trying to specify a buck type converter then the first item that needs to be determined is the minimum inductance. You do this by taking into account the inductor ripple current, the switch on/off times, inductor size, temperature and core saturation. Let’s take, as an example, the following specifications of a Buck type converter to figure the minimum required inductance:  Input Voltage +42V to +54V  Output Voltage +5V  Switching Frequency 75KHz  Max DC Output Current 2.5 Amps  Min DC Output Current 0.5 Amps

Using the maximum input voltage, the following equation will give the minimum of “ontime” that the switch:  Ton MIN=(5+.5)/54x75,000=1.35 uSec

By selecting a value of 1 amp for the inductor ripple current (twice the minimum value of DC output current) then the minimum inductance value can be found:  LMIN=(54-5-0.5)(1.35x10-6)/1= 65.5 uH Therefore, the inductor that needs to be specified for this particular application should have an inductance value of 65.5 uH.

Figure 5.1: Buck 5.3

Boost Inductor design

Typically, the circuit for the Boost type of inductor is very similar to that of the Buck type with one basic difference…the current through the inductor does not flow continuously to the output side. Instead, as you can see in the circuit below, during the time that the switch is on, the output end of the inductor is switched to ground which call for the diode to be reverse biased. So, that means that the entire output current must be supplied by the output capacitor. Now, during the time that the switch is off, the output side of the inductor becomes positive, the diode becomes forward biased which then means a positive output voltage is produced. Of course, the value of the output is determined by the on/off times of the switch. This is the whole crux of determining the inductance to use in a boost type converter. As we did for the Buck inductor we need to determine the following; switch on/off frequency, Input voltage, output voltage and the maximum allowable DC output Current. So if we were to use the same typical specification as before we would find the following values used to determine the inductance level:  Input Voltage +22V to +26V  Output Voltage +5V  Switching Frequency 100KHz

 Max DC Output Current 1.5 Amps  Min DC Output Current Not Specified For this example let’s assume that the maximum inductor ripple current is 12.5% of IL(AVG). The equation would be as follows:  Ton MIN=(24-15+0.5)/100000*24= 3.96 uSec  IL(AVG)=1.05(24+0.5)1.5/15= 2.57 Amps  DIL=2*0.125*2.57=0.643 Amps  L MIN=(15-0.5)(3.96 1 10-6)/0.643=89 uH The minimum inductance level for this inductor needs to be 89 uH.

Figure 5.2: Boost 5.4

Buck/Boost Inductor design

Now, in order to complete the third leg of our Buck/Boost journey, we need to take a look at how to determine minimum inductance values of a Buck-Boost type inductor. These types of converters are very efficient in design and in the use of magnetic components. However, they should only be used when the difference in input voltages are very narrow, say between 15V & 20V . This type of application would be for power that is not subject to wide swings in voltage levels like those coming out of wall plugs or other heavily regulated sources. The same sort of analysis that we used for the Boost inductor is similar to anayzing the needs for Buck-Boost. However, because we’re working with a more complex design due to the dynamics of the switched input current (on/off), we find that the circuit can step a voltage up (boost) or down (buck) but, it also provides a reversal of the polarity. Again, as in the previous examples, we need to pay attention to several key factors but they are limited only to the switch on/off times and the inductor ripple current. The major difference between the two previous examples and the buckboost is the switched values, as in the time during on which allows maximum input current and the time during off which allows minimum input current. So, the equation then becomes;  Lmin=(Ein(max)-0.5)TImin/Il With the following specifications, we can find the minimum required inductance for the Buck-Boost sytle of inductor.  Input Voltage +15V to +20V  Output Voltage -12V  Switching Frequency 40KHz  Max DC Output Current 0.75 Amps

Again, as before, we assume a minimum ripple current of 12.5% of IL(AVG).

The formulae would then be:  TI=12.5/40000 (20+12) = 9.77 uSec  IL(AVG)=1.05*12.5*0.75/(20*40000*9.77*10-6)=1.26 Amps  IL=2*0.125*1.26=0.315 Amps And, finally,  Lmin=(20-0.5)(9.77*10-6)/0.315=605 uH So the minimum inductance value of this component would be 605 uH. Remember, when

Figure 5.3: Buck/Boost 5.5

Pulse Width Modulation(PWM)

Pulse-width modulation uses a square wave whose pulse width is modulated resulting in the variation of the average value of the waveform. If we consider a square waveform f(t) with a low value ymin, a high value ymax and a duty cycle D, the average value of the waveform is given by :

As f(t) is a square wave, its value is ymax for . The above expression then becomes

This

latter

expression

can

be

fairly

and ymin for

simplified

in

many

cases

where

. From this, it is obvious that the average value of the signal () is directly dependent on the duty cycle D.

Figure 5.4 : The PWM model

Figure 5.5: Vin

Figure 5.6: VOUT Table 5.1 : PWM generator chips Manufacture r ST

IC

Normal Use SMPS

Maxim

SG1524 SG3525A MAX038

Atmel

U2352B

TI TI

TL494 UC2638

Comment

May operate at up to 100%duty cycle Signal Generator PWM output only between 15% and 85%.Generate triangle & sine waves too.

PWM Generator Includes integrated current for seep control limiting circuitry for output of portable tools MOSFETs. SMPS Max 90% duty cycle PWM Generator Provides many other features for for motor control DC motor speed control. Note there are many other TI motor control device listed.

5.6 Duty cycle In telecommunications and electronics, the term duty cycle is used to describe the fraction of time that a system is in an "active" state. In particular, it is used in the following contexts:

Figure 5.7 : Duty Cycle The duty cycle D is defined as the ratio between the pulse duration (Ď&#x201E;) and the period (T) of a rectangular waveform.Duty cycle is the proportion of time during which a component, device, or system is operated. Suppose a disk drive operates for 1 second, and is shut off for 99 seconds, then is run for 1 second again, and so on. The drive runs for one out of 100 seconds, or 1/100 of the time, and its duty cycle is therefore 1/100, or 1 percent.

In a periodic phenomenon, the ratio of the duration of the phenomenon in a given period to the period.

Where, D is the so-called duty cycle; Ď&#x201E; is the duration that the function is non-zero;

Τ is the period of the function.

For example, in an ideal pulse train (one having rectangular pulses), the duty cycle is the pulse duration divided by the pulse period. For a pulse train in which the pulse duration is 1 μs and the pulse period is 4 μs, the duty cycle is 0.25. The duty cycle of a square wave is 0.5, or 50%.

In a continuously variable slope delta (CVSD) modulation converter, the mean proportion of binary "1" digits at the converter output in which each "1" indicates a run of a specified number of consecutive bits of the same polarity in the digital output signal.

Some music synthesizers vary the duty cycle of their audio-frequency oscillators to obtain a subtle effect on the tone colors. This technique is known as Pulse-width modulation (PWM).

5.7 The Complete Model

Figure 5.8 : Three phase rectifier with switching.EMI and series LC filter

Table 5.3: THD & Efficiency with duty cycle in single switch boost rectifier with series LC filter D 0.960 0.880 0.848 0.800 0.720 0.640 0.548 0.472 0.316 0.260 0.180 0.122 0.056

THD 0.96 0.98 1.03 1.04 1.14 1.33 1.65 2.01 3.78 3.85 3.37 2.10 2.04

P.F 0.8090 0.8090 0.8090 0.8090 0.8090 1.0000 1.0000 1.0000 0.9500 0.8400 0.7289 0.6100 0.6100

V out 450.00 412.00 640.00 800.00 930.00 900.00 820.00 750.00 635.00 590.00 550.00 510.00 483.00

Ifund 33000 32500 32000 31000 30000 19700 16000 13500 11700 10600 9900 9200 8900

P out 2025 1697.44 4096 6400 8649 8100 6724 5625 4032.25 3481 3025 2601 2332.89

P in 12013.65 11831.63 11649.60 11285.55 12015.00 8865 7200 6075 5001.75 4006.80 3247.25 2350.278 2273.69

Efficiency 0.17 0.14 0.35 0.57 0.72 0.91 0.93 0.93 0.81 0.87 0.93 0.99 0.99

Table 5.4: THD & Efficiency with duty cycle in single switch boost rectifier without series LC filter D 0.8000 0.7580 0.6984 0.6340 0.5680 0.5100 0.4392 0.3840 0.3120 0.2500 0.1848 0.1220

THD 8.10 8.32 7.94 8.72 11.78 14.67 16.73 17.86 22.41 17.60 17.52 16.47

P.F 0.84440 0.8607 0.8952 0.9500 0.9891 0.9990 0.9939 0.9759 0.9510 0.8910 0.8090 0.7070

V out 632 795 928 964 896 821 753 695 645 600 555 520

Ifund 32.33 31.25 30 25.60 20 15 12.70 11 8 9.20 9.80 9

P out 3994.24 6320.25 8611.84 9292.96 8028.16 6740.41 5670.09 4830.25 4160.25 3600.00 3080.25 2704.00

P in 12278.93 12103.59 12085.20 10944.00 8901.9 6743.25 5680.139 4830.705 3423.60 3688.74 3567.69 2863.35

Efficiency 0.3253 0.5222 0.7126 0.8491 0.9018 0.9996 0.9982 0.9999 1.2152 0.9759 0. 8634 0.9344

5.8 The wave shape of input current with only EMI filter for different values of duty cycle are shown below.

Figure 5.9 : Input Current for D=0.068

Figure 5.10 : Input Current for D=0.098

Figure 5.11 : Input Current for D=0.188

Figure 5.12 : Input Current for D=0.43

Figure 5.13 : Input Current for D=0.55

Figure 5.14 : Input Current for D=0.61

Figure 5.15 : Input Current for D=0.67

Figure 5.16 : Input Current for D=0.73

Figure 5.17: Input Current for D=0.8

Figure 5.18 : Input Current for D=0.91 5.9 Different Switching Frequency Switching frequency 0.1k Hz

Figure 5.19: input current & O/p voltage at D=0.70

Figure 5.20: input current & O/p voltage at D=0.50 Figure 5.21: input current & O/p voltage at D=0.25 Switching frequency 0.5k Hz

Figure 5.22 : input current & O/p voltage at D=0.70

Figure 5.23 : input current & O/p voltage at D=0.50

Figure 5.24 : input current & O/p voltage at D=0.25 Switching frequency 1k Hz

Figure 5.25 : input current & O/p voltage at D=0.70

Figure 5.26 : input current & O/p voltage at D=0.50

Figure 5.27 : input current & O/p voltage at D=0.25

Switching frequency 2k Hz

Figure 5.28: input current & O/p voltage at D=0.70

Figure 5.29: input current & O/p voltage at D=0.50

Figure 5.30 : input current & O/p voltage at D=0.25 Switching frequency 3k Hz

Figure 5.31 : input current & O/p voltage at D=0.70

Figure 5.32 : input current & O/p voltage at D=0.50

Figure 5.33 : input current & O/p voltage at D=0.25 Switching frequency 4k Hz

Figure 5.34: input current & O/p voltage at D=0.70

Figure 5.35: input current & O/p voltage at D=0.50

Figure 5.36: input current & O/p voltage at D=0.25 Switching frequency 5k Hz

Figure 5.37: input current & O/p voltage at D=0.70

Figure 5.38: input current & O/p voltage at D=0.50

Figure 5.39 : input current & O/p voltage at D=0.25 Switching frequency 6k Hz

Figure 5.40: input current & O/p voltage at D=0.70

Figure 5.41: input current & O/p voltage at D=0.50

Figure 5.42: input current & O/p voltage at D=0.25 Switching frequency 7k Hz

Figure 5.43 : input current & O/p voltage at D=0.70

Figure 5.44 : input current & O/p voltage at D=0.50

Figure 5.45 : input current & O/p voltage at D=0.25 Switching frequency 8k Hz

Figure 5.46 : input current & O/p voltage at D=0.70

Figure 5.47 : input current & O/p voltage at D=0.50

Figure 5.48 : input current & O/p voltage at D=0.25 TABLE : 5.5 Output Voltage & input Current at different switching Frequency Sw.freq(HZ)

Vref=4v D=0.668

Vref=7v D=0.488

Vref=11v D=0.248

V(avg)

IV1(peak)amp

V(volt)

IV1(amp)

V(volt)

IV1(amp)

0.1k

780

40

920

45

950

28

0.5k

1k

34

830

18

600

11

1k

950

34

700

18

600

12

2k

850

30

760

16

580

10

3k

780

30

700

15

570

10

4k

730

30

680

15

550

10

5k

700

30

650

15

530

10

6k

660

29

620

14

490

10

7k

620

14

610

14

490

10

8k

610

28

600

14

480

10

TABLE : 5.6 Different duty cycle varying Vref as Switching Frequency 4kHz

Vref 0.5 1 2 3 4 5 6 7 8 9 10 11 12 13 13.5

Duty Cycle 0.882 0.852 0.788 0.728 0.668 0.608 0.548 0.488 0.428 0.368 0.308 0.248 0.188 0.128 0.098

Input voltage output voltage at different input voltage (Switching frequency Vref 4V)

Figure 5.49 : Input voltage 10v

Figure 5.50 : Input voltage 17v

Figure 5.51 : Input voltage 20v

Figure 5.52 : Input voltage 30v

Figure 5.53 : Input voltage 40v

Figure 5.54 : Input voltage 50v

Figure 5.55 : Input voltage 80v

Figure 5.56 : Input voltage 100v

Figure 5.57 : Input voltage 150v

Figure 5.58 : Input voltage 200v

Figure 5.59 : Input voltage 220v

Figure 5.60 : Input voltage250v

Figure 5.61 : Input voltage 280 v

Figure 5.62 : Input voltage 300 v Table 5.7 : Input voltage , output voltage and input current Vin(V)(peak)

Vout(V)avg

I in(Amp)(peak)

10

30

1

17

48

1.9

20

65

2

30

94

3.15

40

130

4

50

162

5.8

80

250

8.5

100

310

10

150

430

15

200

550

20

220

590

21

250

650

23.5

280

700

25

300

730

28

Chapter 6 CONCLUSIONS & SUGGESTIONS

6.1 Introduction In a three phase rectifier circuit the input currents are non sinusoidal in nature & contains harmonics. The power infrastructure has to carry these currents & should be oversized to accommodate the flow of harmonics. Harmonics produce heat loss, lower efficiency, input ac mains voltage distortion, lower power conversion reliability, excitation of undesirable system resonances & increased VA rating of the equipments. This is why harmonic reduction is one of the main concerns. An easy technique to smooth out the input current is to use boost rectifier where the added inductor shape the input current from not being pulsating. But this leads to lessen the value of the output voltage. That is why active switching had been introduced. Different authors had already studied rectifiers with active switching & EMI filter. Even with the incorporation of EMI filter the THD value does not go below acceptable range. So in our work we have studied the effect of another series LC resonating filter to get an acceptable THD limit of the three phase rectifier input current.

6.2 Conclusions First of all we had studied the normal rectifier circuit with & without capacitor. Without capacitor the THD value of a normal three phase rectifier that we have designed is 22%. But when the capacitor was included at the output to filter the voltage ripple the THD value rise to 81%. Another drawback of this circuit is that it does not have voltage control option.

Then we studied the boost rectifier circuit & observed the improvement over normal rectifier.

For boost action inductors are placed in the input side. The inductive behavior improves the input current shape from pulsating to be uniformly varying in nature. As we increase the value of the inductor the wave shape improves significantly but the output voltage decreases & in this circuit voltage regulation is also not possible.

Next boost rectifier with active switching has been studied. The switching circuit has PWM module for regulation of the output voltage. In this circuit the input current is DCM in nature & contains high frequency component. Though the average input current is sinusoidal in nature, this type of current would radiate interference in the nearby electronics equipments will hamper their normal operation.

Electro Magnetic Interference (EMI) filter is added in a boost rectifier with PWM module.

Behavior of this module has been studied. It was found that the THD value has reduced to 17%. The input current wave shape is not purely sinusoidal. It contains low frequency components.

To suppress the low frequency component from the input current another series resonating filter has been introduced, which resonate at the supply frequency. With this added resonating filter the THD value has reduced to less than 4%. The wave shape is nearly pure sinusoid & voltage could be regulated from 400V to 1KV. But if the efficiency is also the concern then the choice of a certain range of duty cycle to operate the module is necessary.

Our best operating point is at the range of duty cycle 0.47 to 0.64. At this range the power factor is unity & output voltage can be regulated from 750V to 900V & efficiency is 91% to 93% & the THD value is less than 2%.

6.3 Suggestions for future work

In this work a resonant filter is used at the input side to filter low frequency components. The resonant filter resonates at line frequency (50Hz). As a result the filtering components are very large in values. Any other method like series parallel combination of LC filter to reduce the component values may be studied in future to see whether the component values can be lowered.

Only simulation is performed in this study. The work was not implemented practically. So another suggestion for future work of this thesis would be to implement the practical circuit.

The practical circuit would require implementing two parts.

(1) Power circuit design (2) Boost switching stage Power circuit had been already designed in normal rectifiers. So the concern is to design the switching stage, which has four more stages to implement; (a) (b) (c) (d)

Control circuit, Base drive, Base isolation & Voltage controller.

The design of control circuit, base drive & base isolation stages are already given in this study. But the voltage control circuit has to be implemented as required by loads. Future work may include designing voltage control circuit.

APPENDIX A.

MC33260

A.1 GreenLineTM Compact Power Factor Controller The MC33260 is a controller for Power Factor Correction pre-converters meeting international standard requirements in electronic ballast and off−line power conversion applications. Designed to drive a free frequency discontinuous mode, it can also be synchronized and in any case, it features very effective protections that ensure a safe and reliable operation. This circuit is also optimized to offer extremely compact and cost effective PFC solutions. While it requires a minimum number of external components, the MC33260 can control the follower boost operation that is an innovative mode allowing a drastic size reduction of both the inductor and the power switch. Ultimately, the solution system cost is significantly lowered. Also able to function in a traditional way (constant output voltage regulation level), any intermediary solutions can be easily implemented. This flexibility makes it ideal to optimally cope with a wide range of applications.

A.2

General Features

Standard Constant Output Voltage or “Follower Boost” Mode Switch Mode Operation: Voltage Mode Latching PWM for Cycle−by−Cycle On−Time Control Constant On−Time Operation That Saves the Use of an Extra Multiplier Totem Pole Output Gate Drive Under voltage Lockout with Hysteresis

Low Startup and Operating Current Improved Regulation Block Dynamic Behavior Synchronization Capability Internally Trimmed Reference Current Source Pb−Free Packages are Available A.3

Safety Features

Overvoltage Protection: Output Overvoltage Detection Under voltage Protection: Protection Against Open Loop Effective Zero Current Detection Accurate and Adjustable Maximum On−Time Limitation Over current Protection ESD Protection on Each Pin A.4

Pin Connections

A.5

Marking Diagrams

Figure A.1: Block Diagram

Figure A.2: Typical Waveforms A.6

Package Dimensions

B.

NCP1650

B.1

Power Factor Controller

The NCP1650 is an active, power factor correction controller that can operate over a wide range of input voltages, and output power levels. It is designed to operate on 50/60 Hz power systems. This controller offers several different protection methods to assure safe, reliable operation under any conditions. The PWM is a fixed frequency, average current mode controller with a wide complement of features. These features allow for both flexibility as well as precision in itâ&#x20AC;&#x2122;s application to a circuit. Critical components of the internal circuitry are designed for high accuracy, which allows for precise power and current limiting, therefore minimizing the amount of overdesign necessary for the power stage components.

The NCP1650 is designed with a true power limiting circuit that will maintain excellent power factor even in constant power mode. It also contains features that allow for fast transient response to changing load currents and line voltages.

B.2

Features

• Fixed Frequency Operation • Average Current Mode PWM • Continuous or Discontinuous Mode Operation • Fast Line/Load Transient Compensation • True Power Limiting Circuit • High Accuracy Multipliers • Undervoltage Lockout • Overvoltage Limiting Comparator • Brown Out Protection • Ramp Compensation Does Not Affect Oscillator Accuracy • Operation from 25 to 250 kHz • Pb−Free Package is Available* B.3

Typical Applications

• Server Power Converters • Front End for Distributed Power Systems B.4

Marking Diagram

B.5

Pin Connections

88

Figure B.1: Simplified Block Diagram

Figure B.2: Timing Diagram

B.6

Package Dimensions

References [1] Richard Red1 ELFI S.A. Derrey-la-Cabuche Power-Factor Correction in Bridge and Voltage-Doubler Rectifier Circuits with Inductors and Capacitors.CH-1756 Onnens (FR) Switzerland [2] J. M. Bourgeois, â&#x20AC;&#x153;Circuits For Power Factor Correction With Regards To Mains

Filtering” [3] Wei, H.,"Single-Stage Single-Switch Power FactorCorrection Circuits:Analysis, Design and Implementation," Ph.D. dissertation, University of Cenbal Florida, 1999 [4] Jovanovic, M.M.; Crow, DE,"Merits and Limitations of Full-BridgeRectifier with LC Filter in Meeting IEC 1000-3- 2 Harmonic-Limit Specifcations," IEEE Transactions on Industry Apdications,Volume: 33 Issue:2, March-April1997,Pageis): 551-557.

[5] Vomerian. V. Ridlev. R.B. "A Simde Scheme for Unim Power Factor 'Recfifiction/ orHigh FrequencyAb Buses>'' IEEE Transactions on Power Electronics, Volume: 5 Issue: I , Jan. 1990,Page@):77-87.

[6] Sharifipour, B.; Huang, J.S.; Liao, P.; Huber, L.; Jovanovic,M.M.,"Manufacrtrringand Cost Analysis of Power-Factor- CorrectionCircuits," AppliedPower Electronics Conference and Exposition, 1998. AF'EC '98. Conference Proceedings 1998,Thirteenth Annual, Volume: I, 1998,Page(s): 490-494 v0l.l.

[7] Q.Huang & F.C. Lee, “Characterization & Control of Three-Phase Boost Rectifier at Light Load”, Proceedings of the Virginia Power Electronics center seminar 1996, pp.29-34.

[8] http://en.wikipedia.org/wiki/Pulse-width_modulation [9] R. Liu, "Analysis and Design of High-Order Resonant Converters and A Unified.

Approach to Power Factor Correction," Ph.D. Thesis, Univ. of Illinois, Chicago, August'91 [10] S. Cuk, "Modeling, Analysis, And Design of Switching Converters," Ph.D. Thesis, California Inst, of Tech. 1977 [11] Peter Kornetzky, Huai Wei, Guangyong Zhu and Issa Batarseh, "Asingle-Switch Ac/Dc Converter with Power Factor Correction,"Electronics Letters, Dec. 1997, vol. 33, no. 25, pp. 2084-2085.

91 [12] J. Qian, I. Batarseh and M. Ehsani, "Analysis and Design of A Clamp-Mode Isolated Zero-Voltage Switching Boost Converter," IEEE APEC'95 Proc, pp. 1201-1206.

[13] R. Redi, "Reducing Distortion in Boost Rectifiers with Automatic Control," IEEE APEC'97 Proc, pp. 74-80.

[14] Y. Jiang and F. C. Lee, "Single Stage Single- Phase Parallel Power Factor Correction Scheme," IEEE PESC'94 Proc, pp. 1145-1151.

[15] R. Waston, G. C. Hua, and F. C. Lee,"Characterization of and Active Clamp FlybackTopology for Power Factor Correction Applications," IEEE APEC'94 Proc, pp. 412418.

[16] R.Erickson, M.Madigan, and S.Singer, "Design of a Simple High- Power Factor Rectifier Based on the Flyback Converter," IEEE APEC'90 Proc, pp. 792-801.

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