Power Electronics Handbook 2021 by WTWH Media LLC

Handling the 48-to-12-V stepdown Page 12 The changing maze of power supply standards Page 32 Selecting dc-link capacitors for inverters Page 36

FEBRUARY 2021

Power Electronics Handbook

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Battery Clips, Contacts & Holders

Radar: From Wars to Baseball

From our Blog Series — Radar: From Wars to Baseball We’re perhaps most familiar with the use of radar guns by the police for catching speeders — a practice started shortly after World War II. It wasn’t long before Radio Detection and Ranging — “radar” was used in baseball to measure the speed of each pitch. Radar, like so many other high tech devices today, require precision electronic components for their successful operations. You’ll find many products inside the radar gun clocking that pitch or car such as Keystone’s featured Battery Clips, Contacts & Holders Other products you’ll find in the radar gun include: • Anti-Vibration Grommets • Key Pad Dome Switches • LED Spacers and Lens Caps • Pins, Plugs, Jacks & Sockets

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Battery breakthrough fatigue For

some cheap thrills, try Googling the words “battery breakthrough.” You’ll be presented with page after page of articles breathlessly chronicling one research result after another in energy storage. But you might wonder why, with so many earth-shattering developments, electric vehicles can’t travel 1,000 miles before needing a recharge. The reason: It doesn’t take much to be a breakthrough in batteries these days, at least not in the eyes of journalists. We did a quick review of the top “battery breakthrough” search results to figure out why EVs in our neighborhood are forced to charge up every night. What we found was that a lot of the work reported in excited tones is over hyped. The typical “battery breakthrough” isn’t a battery at all. It more likely takes the form of goop in a flask sitting on a lab bench somewhere. That said, researchers quoted in write-ups of their work tend to be circumspect about what they’ve achieved. Unlike the gushing of reporters, individual researchers are more likely to characterize their accomplishments as building blocks or promising strides on what’s likely to be a long road to commercialization. For example, consider the “breakthrough” at Nanotech Energy, a startup that hopes to produce a battery charging “18 times faster than anything currently on the market” within the year. Perhaps, but all it has made so far are graphene inks. Then there are the electrolytes coming out of the Lawrence Berkeley National Lab that suppress dendrite growth on battery anodes before it can cause the battery to fail. The electrolyte is flexible enough to be a laminate between the anode and the battery separator on rolls of lithium foils--someday. But that’s down the road a bit. Also in this category is Sila Nanotechnologies. It has come up with a “nanocomposite” of silicon and other

materials to replace the graphite in electrodes. The company says its product will boost battery capacities by 20 to 40%. But Sila doesn’t actually make batteries. It hopes to get its technology into the hands of companies that do, in fact, make batteries in the next year. A similar development comes from scientists at Washington State University. Researchers there were able to prevent dendrites from forming on electrodes—and eventually shorting out the battery—by adding a few key chemicals to the cathode and electrolyte to form a protective layer on the surface of a lithium-metal anode. The team figures the process can be integrated into existing manufacturing procedures. But no word yet of any battery maker actually doing so. Reading between the lines, many advances touted as breakthroughs sound like maybe-somebody-can-use-this developments. A minority involve real batteries. One of the few in this category is QuantumScape. The company has released test data on its solid-electrolyte cells indicating batteries based on them could charge 80% in just 15 minutes. But the QuantumScape battery cells are just one layer of a much larger cell structure that will make up real batteries. It remains to be seen whether QuantumScape can scale up and produce entire battery packs at industrial scale. If things go well, Volkswagen says it hopes to use the batteries in its 2024 or 2025 vehicles. So though QuantumScape is well beyond the goop-in-a-flask stage, its technology is still about five years away from commercialization. That time frame is something to keep in mind the next time you see a headline announcing “breakthrough” battery chemistry lab work.

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CONTENTS POWER ELECTRONICS HANDBOOK FEBRUARY 2021

02 06

Battery breakthrough fatigue

No zapping! How to design safe and reliable wireless charging systems

Super-efficient power converters help hovering drones take flight

The changing maze of power supply standards

SMART USE OF PROTECTION COMPONENTS CAN HELP GIVE CONSUMERS A TROUBLE-FREE CHARGE.

12 16 20

Handling the 48-to-12-V stepdown A MULTIPHASE, INTERLEAVED BUCK CONVERTER CAN REDUCE BUS VOLTAGE DOWN TO SOMETHING USEFUL IN DATA CENTER SERVER RACKS.

A FEW BASIC PARAMETERS CAN BE USEFUL FOR SELECTING POWER SUPPLIES THAT MUST WORK DEPENDABLY IN SPECIFIC APPLICATION SCENARIOS.

Maximizing the performance of SiC through gate drive techniques

How to boost output hold-up time in power supplies

MANY SUPPLIES HAVE OUTPUTS THAT DON’T STAY UP LONG ENOUGH IN THE EVENT OF A POWER LOSS. HERE ARE SOME WAYS TO REMEDY THE PROBLEM. eeworldonline.com

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Selecting dc-link Capacitors for inverters ONE KEY FACTOR: DETERMINING THE NUANCES OF HOW CAPACITORS HANDLE EXPECTED RIPPLE CURRENTS.

Picking reliable power supplies

NEW GATE DRIVERS SIMPLIFY THE TASK OF DEPLOYING SILICON-CARBIDE FETS TO SHRINK THE SIZE AND BOOST THE EFFICIENCY OF POWER INVERTERS.

BREXIT ALTERED THE LANDSCAPE OF STANDARDS FOR POWER SUPPLY PRODUCTS. HERE’S AN UPDATE ON THE LATEST REVISIONS.

The advantages of Generation-Four SiC FETs

NEW SILICON-CARBIDE FETS PERFORM BETTER THAN PREVIOUS VERSIONS AND CAN REPLACE SILICON MOSFETS WITH RELATIVE EASE.

Fundamentals of fast chargers for phones and notebook PCs

OVERLAPPING PROTOCOLS AND COMPETING STANDARDS HAVE CREATED A CONFUSING LANDSCAPE FOR USB CHARGERS. HERE ARE THE IMPORTANT DIFFERENCES AND SIMILARITIES OF MOST INTEREST TO DESIGNERS.

2 • 2021

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No zapping! How to design safe and reliable wireless charging systems SMART USE OF PROTECTION COMPONENTS CAN HELP GIVE CONSUMERS A TROUBLE-FREE CHARGE.

The

convenience that wireless charging systems provide have made them increasingly popular for use with consumer goods and appliances. But the development of wireless charging devices creates a number of challenges for designers. Most wireless applications are characterized by limited space for circuitry and cost considerations. So designers must minimize component counts without compromising quality. Simultaneously, designs must also protect against failures related to overloads and transients. And energy efficiency is usually a priority as well. Also, products must comply with international standards for safety, surge and transient protection. Thus it is useful

TODD PHILLIPS

LITTELFUSE INC.

to understand the methods available for protection, efficient control, and safety sensing of wireless chargers. A wireless charging system consists of three elements; a power adapter, a charging cable, and a charging pad. The power adapter converts the ac voltage from the power line into a dc voltage. The charging cable sends power from the power adapter to the wireless charging pad. The charging pad then transmits power wirelessly to the device-to be-charged. The power adapter must contend with any overload and transient conditions presented by the ac mains. Circuit protection generally is at the input stage. Common electrical hazards include; induced lightning voltage surges, switching surges, electrostatic discharge,

and overload fault currents. Low-power power adapters use a variety of overvoltage protection schemes. MOVs are often the choice for power adapters where long-term reliability is a priority. Fuses are the primary choice for overcurrent protection, especially for higherpower adapters. Designers have a wide range of options when deciding on the fuse form factor. The main choices are cartridge fuses, thru-hole fuses, or surface-mount fuses. Surface-mount fuses typically take up the least amount of PCB real estate. Regardless of fuse form factor, the fuse must have voltage and current-interrupting ratings sufficient to do the job. Designers frequently consider a time-lag fuses to

PRODUCT SAFETY STANDARDS STANDARD

TITLE

GENERAL SCOPE

REGION

USB-IF

Universal Serial Bus specification

Supports advancement and adoption of Universal Serial Bus technology

Global

IEC 60950-1

Information technology equipment - safety

Applicable to mains-powered or battery-powered information technology equipment, with a rated voltage not exceeding 600 V

Global

IEC 62368-1 (IEC 60950)

Audio/video, information and communication Safety of equipment within the field of audio, video, technology equipment - Part 1: safety information and communication technology (rated voltage requirements not exceeding 600 V)

IEC 61000-4-2

Testing - Electrostatic Discharge (ESD)

This standard is made to check the capability of equipment to survive repetitive electrical fast transients and bursts

Global

IEC 61000-4-4

Electrical fast transient/burst immunity test

Evaluating the immunity of equipment when subjected to electrical fast transient/bursts on supply, signal, control, and earth ports

Global

IEC 61000-4-5

Surge immunity

Evaluate the immunity of equipment when subjected to surges

Global

Qi Wireless Power Transfer System

Open interface standard defining wireless power transfer using inductive charging

Global

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Global

INTERNATIONAL STANDARDS FOR PRODUCT SAFETY AND COMPLIANCE, USB COMMUNICATION STANDARDS, AND THE QI WIRELESS POWER TRANSFER STANDARD.

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WIRELESS CHARGING PROTECTION WHERE TO FIND PROTECTION COMPONENTS Charging pad

Charging cable

Power adapter Fuse TVS Diode N-channel MOSFET Barrier rectifier diode Digital Temperature Indicator (setP™ ) *

Digital Temperature Indicator (setP™ ) *

Fuse TVS Diode, Diode Array Digital Temperature Indicator (setP™ ) *

WHERE CIRCUIT PROTECTION COMPONENTS ARE TYPICALLY DEPLOYED IN A WIRELESS CHARGING SYSTEM. Acronyms: TVS: transient voltage suppressor Protect

avoid nuisance tripping from overvoltage events. Each fuse type responds differently to overloads. In addition, designers trying to maximize energy efﬁciency should evaluate the fuse watts-loss rating. Designers can maximize charger efﬁciency by selecting MOSFETs with low on-state resistance, low gate charge, and a high dv/dt rating to reduce switching loss and obtain faster switch transition times. MOSFETs with low on-state resistance and high dv/dt parameters can switch at higher

Sense

* For use with a USB Type C adapter

frequencies, enabling a more efﬁcient switchmode supply circuit topology. For the same reasons, designers should use MOSFETs with internal soft-recovery diodes which reduce turn-off transients and reduce electromagnetic emissions (EMI). In many supply topologies, a step-down transformer reduces an ac voltage and Schottky diodes then rectify the signal back to dc. Schottky diodes with low forward-voltage-drop and which can operate at high frequencies are suggested for this section of the design.

TVS DIODES Bi-directional

Cathode

Anode Uni-directional

TVS DIODE CONFIGURATIONS FOR UNIDIRECTIONAL AND BI-DIRECTIONAL TRANSIENT PROTECTION.

USB POWER PROTECTION AC mains

Input protection, rectifier, and filter

Highfrequency converter and clamp

3 Step-down transformer

Output protection, rectifier, and filter

Technology

4 USB Type C controller and power switch

USB Type C (Port protection)

To peripherals

Fuse

N-channel MOSFET TVS diode

Power adapter

Legend:

Feedback and PWM oscillator

Schottky diode Digital Temperature Indicator1

1The

USB Plug

Power Line Signal Line

digital temperature indicator (setP™) solution is recommended for USB Type C port protection. Use Low Rho SMD Series for USB Type-A and USB Type-B

A TYPICAL BLOCK DIAGRAM FOR THE USB POWER ADAPTER AND CHARGING CABLE WITH TYPICAL CIRCUIT PROTECTION MEASURES INCLUDED.

Charging cable

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Resistance, (Ω)

POLYSWITCH setP R vs T

Temperature, °C

Transient voltages on the input of the supply can find their way to the output rectifiers. Some of these transients may be large enough to damage the powersemiconductor devices. Transient voltage suppression (TVS) diodes can provide a level of protection. TVS diodes can respond to a transient extremely quickly, typically in under one picosecond. They also have low clamping voltages to protect sensitive electronic circuits. Designers can select either uni-directional or bidirectional configurations of a TVS diode. The USB Type-C protocol allows up to 100 W of charging to speed the re-charging process. This is a substantial increase in 1available power over earlier USB standards. This high-level power protocol uses USB Type-C connectors which have a 0.5-mm pitch, five times smaller than USB Type-A connectors. The application of more power in less space increases the risk that dust and dirt Littelfuse, Inc. © 2020

RESISTANCE VS TEMPERATURE CHARACTERISTIC CURVES FOR POLYSWITCH SETP TEMPERATURE INDICATORS.

will create a resistive fault between pins on the connector and create a hot spot. Designers should consider using a digital temperature indicator, like the PolySwitch setP, to detect over-temperatures. The setP temperature indicator measures less than a tenth-of-an-inch square and exhibits a rapid rise in resistance as temperatures increase from about 80 to 100°C. It is designed to be used in the Configuration Channel (CC) line of the Type-C connector to detect over-temperature events and help protect the circuit. The setP temperature indicator complies with the USB Type-C standard for monitoring the temperature of USB Type-C connectors. Details on the circuit configuration for this protection scheme are in the USB Type-C cable and connector specification.

TYPICAL WIRELESS CHARGING PAD Technology

CP DC input from adapter

Bridge inverter

DC input

1a DC-DC converter

USB input

C-C signal line for USB-C connector

MCU

C Charging Pad Legend:

Power Line Signal Line

High frequency gate driver

Wireless power transmitter IC

To mobile device

Fuse TVS Diode Digital Temperature Indicator Diode Array

Acronyms: TVS: transient voltage suppressor MCU: microcontroller unit ESD: electrostatic discharge suppressor Notes: 1a: DC jack or 1b: The USB-C port; digital temperature indicator (setP™) solution is suitable for USB Type C port protection (generally, only one DC-input option is implemented in a unit).

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POWER ELECTRONICS HANDBOOK PROTECTING THE WIRELESS CHARGING PAD Power for the wireless charging pad comes from either a proprietary dc input or a USB port. Designers should protect the dc input circuit from both overloads and transients. Regardless of the type of power input used, protection is highly recommended. On the dc input circuit, designers should consider overload protection via fastacting fuses. Small surface-mount fuses rated for the proper dc voltage are suited for this purpose. For transient protection, surface-mount TVS diodes are available which can provide up to ± 30 kV of ESD protection and 1,500 W of peak transient power absorption. Low clamping voltages, common for most TVS diodes, help avoid stressing downstream circuit components when a transient strikes. Of course, wireless chargers must comply with international standards. The standards deﬁne minimum safety requirements and spell out tests that evaluate various electrical hazards such as ESD, electrical fast-transients, and surge levels. Wireless chargers using USB communication must ensure interoperability according to the Universal Serial Bus (USB) standard. Designers should also be familiar with the Qi wireless charging protocol that deﬁnes how energy transfers to product batteries. Circuit protection can help ensure a positive end-user experience. In that regard, manufacturers of protection components can make available application engineers early in the design cycle who can save substantial development time and reduce design revisions.

REFERENCES UNIVERSAL SERIAL BUS TYPE-C CABLE AND CONNECTOR SPECIFICATION. REVISION 2.0. AUGUST 2019. USB IMPLEMENTERS FORUM (USB-IF), INC. HTTPS://USB. ORG/DOCUMENT-LIBRARY/USB-TYPE-CR-CABLE-ANDCONNECTOR-SPECIFICATION-REVISION-20-AUGUST-2019 LITTELFUSE CIRCUIT PROTECTION SELECTION GUIDE HTTPS://INFO.LITTELFUSE.COM/CIRCUITPROTECTION-PRODUCT-SELECTION-GUIDE?UTM_ SOURCE=ARTICLE&UTM_MEDIUM=DESIGNWORLD-POWERHANDBOOK&UTM_CAMPAIGN=CES-MOB LITTELFUSE POLYSWITCH SETP DESIGN GUIDE HTTPS:// INFO.LITTELFUSE.COM/SETP-DESIGN-GUIDE-LF?UTM_ SOURCE=ARTICLE&UTM_MEDIUM=DESIGNWORLD-POWERHANDBOOK&UTM_CAMPAIGN=CES-MOB

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POWER ELECTRONICS HANDBOOK

Handling the 48-to-12-V stepdown A MULTIPHASE, INTERLEAVED BUCK CONVERTER CAN REDUCE BUS VOLTAGE DOWN TO SOMETHING USEFUL IN DATA CENTER SERVER RACKS. BRAD XIAO, NAZZARENO (RENO) ROSSETTI MAXIM INTEGRATED 12

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2 • 2021

Hyperscale

data centers that dissipate hundreds of megawatts demand efﬁcient power delivery beyond the ability of the traditional 12-V power supplies on old server motherboards. A data center on 12 V has a power usage effectiveness (PUE) of 1.1 at best. It wastes 10% of its power (tens of megawatts) on cooling and overhead. That wasted power could comfortably supply electricity to a small city with about 10,000 homes. eeworldonline.com

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DATA CENTER POWER

The move from 12 to 48-V power brings several advantages. The server rack dissipates heat more efﬁciently, costs less, and can be smaller as a consequence of the reduced current (4x), copper losses (16x), and connector/bus bar sizing. In the power management architecture of a modern data eeworldonline.com

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center, different strategies are adopted to power different classes of loads like processors, memories, and peripherals. Peripherals such as PCIE and HDDs require a 12-V supply rail. One way to get the required 12 V is through use of a multiphase, interleaved, coupled-inductorbased 48-to-12-V step-down converter. This 2 • 2021

converter design is compact, efﬁcient, and cost-effective. The best strategy to power low-voltage (< 1.8 V), high-current loads like memories and microprocessors is to reduce the 48 V down to an unregulated 12-V rail. An open-loop, unregulated 4:1 switched-tank converter (STC) DESIGN WORLD — EE NETWORK

POWER ELECTRONICS HANDBOOK ANATOMY OF 48-V SERVER POWER TYPICAL 48-V SERVER ARCHITECTURE TODAY POWERS LOW-VOLTAGE (< 1.8 V), HIGH-CURRENT LOADS LIKE MEMORIES AND MICROPROCESSORS VIA AN UNREGULATED 12-V RAIL. AN OPEN-LOOP, UNREGULATED 4:1 SWITCHED TANK CONVERTER (STC) PRODUCES THE UNREGULATED 12 V. (STCS USE LC RESONANT TANKS TO PARTIALLY REPLACE FLYING—I.E. FLOATING WITH RESPECT TO GROUND--CAPACITORS FOR ENERGY TRANSFER. HIGH-EFFICIENCY 12-V VOLTAGE-REGULATING CONVERTERS POWER THE VARIOUS LOADS.

topology enables soft-switching operation with peak efﬁciency in the 98% to 99% range. From here, high-efﬁciency 12-V voltage regulatorconverters power the various loads. Peripherals like PCIE and HDDs require a 12-V supply rail and currents up to 100 A (1.2 kW). In this case, a 48-to-12-V regulated, interleaved, multiphase dc-dc converter can do the job. Use of a four-phase interleaved dc-dc

converter reduces ripple current and, hence, ripple voltage, compared to that of a single-phase converter. Lowering output current ripple and voltage ripple means fewer capacitors on the output, resulting in a smaller BOM. The four-phase architecture also requires fewer input capacitors. The total input current is the sum of the four out-of-phase currents. Here, spreading the total input current over time reduces the total RMS

STEP-DOWN REGULATION

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2 • 2021

THE 8-TO-60-V MAX15157B FIXEDFREQUENCY, ACCURATE CURRENT REPORT, CURRENT-MODE PWM CONTROLLER DRIVES TWO POWER MOSFETS IN THIS BUCK CONFIGURATION, ALLOWING OPERATION AS A 48-TO12-V STEP-DOWN REGULATOR.

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DATA CENTER POWER FOUR-PHASE DC-DC EFFICIENCY (48V IN , 12V OUT )

A FOUR-PHASE, COUPLED-INDUCTOR CONFIGURATION YIELDS AN OUTSTANDING 97.9% PEAK AND 97.43% FULL-LOAD EFFICIENCY (12 V, 100 A, 1.2 KW) WITH THE INTERLEAVED BUCK CONTROLLER.

THE 48-TO-12-V PCB WITH A COUPLED INDUCTOR. THE DESIGN EASILY FITS ON A MOTHERBOARD. IT FITS IN A QUARTER-BRICK FOOTPRINT USING ONE PCB SIDE, IN AN EIGHTH-BRICK FOOTPRINT ON TWO PCB SIDES.

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value of the input current compared to that of a single-phase setup. This spreading makes possible a smaller input current-ripple filter. Another great benefit of a four-phase converter is a fast transient response and lower voltage overshoot/undershoot during load steps. Lower overshoots, in turn, allow a tighter voltage rail powering the electronic load with less headroom needed for transient droop, resulting in less power dissipated at the load. Additional efficiencies can be had through use of a coupled inductor in which the inductor coils wind around a common magnetic core. Here, the winding orientation is such that the magnetic flux cancels out, resulting in less than a quarter of the ripple current compared to that of four uncoupled inductors. Extremely low ripple current yields the smallest BOM and highest efficiency. An example of a device that makes possible efficient 48-to-12-V conversion is the 8-to-60-V MAX15157B fixed-frequency, current-mode PWM controller. It drives two power MOSFETs in a buck configuration, synthesizing a 48-to12-V step-down regulator. Four ICs--in conjunction with external discrete MOSFETs, capacitors, and a single coupled inductor--are interleaved for multi-phase operation. The switching frequency is controlled either through an external resistor setting the internal oscillator frequency or by synchronizing the regulator to an external clock. The device

2 • 2021

is designed to support switching frequencies ranging from 120 kHz to 1 MHz. Each IC is available in a 5x5-mm, 32-pin TQFN package and supports a -40 to +125°C junction temperature range. In a four-phase, coupled-inductor configuration, this topology yields an outstanding 97.9% peak and 97.43% full-load efficiency (12 V, 100 A, 1.2 kW). The 48-to-12-V PCB layout with a coupled inductor can easily sit on the data server motherboard. It fits in a quarter-brick size on one side of a PCB and in an eighth-brick form factor on two PCB sides. In a nutshell, high levels of efficiency available through use of such 48-to-12-V conversion techniques offer a path toward power usage effectiveness approaching unity in a notso-distant future.

REFERENCES HTTPS:// MAXIM SUPPORT CENTER MAXIMINTEGRATEDSUPPORT.FORCE.COM/ SUPPORT/S/ DESIGN SOLUTIONS HTTPS://WWW. MAXIMINTEGRATED.COM/EN/DESIGN/TECHNICALDOCUMENTS/DESIGN-SOLUTIONS.HTML

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Picking reliable power supplies A FEW BASIC PARAMETERS CAN BE USEFUL FOR SELECTING POWER SUPPLIES THAT MUST WORK DEPENDABLY IN SPECIFIC APPLICATION SCENARIOS. ALEX KARAPETIAN ACOPIAN POWER SUPPLIES

Electronic

devices require a reliable source that can supply power at all times. But power supplies are available in a wide range of voltage and current ratings. Engineers tasked with selecting an appropriate power supply must resolve numerous questions to determine which supply suits an application. A good place to start with this analysis is to look at two main power supply design topologies: switching and linear regulated. Switching power supplies regulate the output voltage using a high-frequency switching technique that employs pulse-width modulation and feedback. Power passes from the input to the output through a switch which is actuated until the desired voltage is reached. When the output voltage reaches the predetermined value, the switch element is turned off, and no power is consumed. Generally speaking, switching power supplies are smaller, lighter, and more efficient than linear power supplies. For example, a 250-W linear power supply would take up a volume of 600 in3 and weigh 26 lb while a comparable ac-dc switching supply would require 60 in3 and weigh 2 lb. Switching supplies can either step-up or step-down the input voltage to get the desired output voltage. Applications where a switching power supply may be preferable include those where small size, low weight, and high energy efficiency are important. They include high power/high current uses, portable equipment, control systems, dc motors, aviation and marine applications, network equipment, electrolysis and waste treatment operations. Linear-regulated power supplies earned their name from use of linear (non-switching) techniques to regulate the supply voltage output. A linearregulated power supply generates an output voltage by first converting the high-voltage ac into lower-voltage ac via a transformer, then converting the transformer output into an unregulated dc voltage via a rectifier and capacitor filters. An error amplifier compares the reference to the output and the resulting signal is used to ensure the output remains on the required voltage. The closed-loop design ensures the supply output stays at the nominal voltage despite changes in supply line or load values.

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AN EXAMPLE OF A REDUNDANT POWER SUPPLY PACKAGE. THE ACOPIAN R24W9 IS A RACKMOUNTABLE SUPPLY THAT PUTS A NOMINAL 24 V AND 24 A. IT IS A SWITCHING SUPPLY, THOUGH REDUNDANT SUPPLIES EMPLOYING LINEAR-REGULATION TOPOLOGIES ARE ALSO AVAILABLE. TYPICAL OPTIONAL FEATURES FOR SUCH REDUNDANT SUPPLIES INCLUDE ALARMS WHEN AN OUTPUT IS BELOW NORMAL, SEPARATE AMMETERS FOR EACH OUTPUT, AND SEPARATE ALARM CONTACTS FOR EACH SUPPLY.

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DEPENDABLE SUPPLIES Linear power supplies typically step-down an input voltage to produce a lower output voltage. Though it is larger, heavier, and not as efficient as a comparable switching supply, the linear unit has beneficial attributes that the switcher supply cannot match. For example, the linear supply has no discrete time clocking or switching action. Consequently, the output is virtually noise and ripple free. Typical applications of linear regulated power supplies include telecommunications equipment, medical equipment, laboratory test equipment, low-noise amplifiers, advanced signal processing and data acquisition systems (sensors, multiplexers, A/D converters and sample and hold circuits), and precision measurement devices.

FAILURE PROOF Systems where reliability is essential and a sudden loss of power would lead to disaster need two bullet-proof power sources. One way to keep dc power going no matter what is to use a redundant power supply system. In a redundant arrangement, more than one power supply feeds a single voltage rail. If one power supply fails, the other continues to entirely power that rail. Furthermore, a separate feed to each power supply input helps stem failure on the primary side. Redundant power entails more than simply paralleling the outputs of multiple power supplies. (In general, power supplies may not be built to work in parallel with other supplies. Simple parallel power connections can sometimes damage the paralleled supplies.) Instead, redundant power packages contain two identical power supplies with their outputs interconnected through a diode switching arrangement. The switching arrangement will detect any fault condition, isolate it from the system output, and pass only the output of the good supply with no interruption of output power during the transition. Redundant power systems come with various features including over- and undervoltage, surge protection, isolation diodes, alarms and remote voltage sensing. Power supplies that can be mounted on a DIN rail are frequently utilized in control panels and electrical cabinets for industrial control, building control, instrumentation and automation applications. Control systems typically have functional modules clipped to DIN rails for operational flexibility and easy access. These DIN-rail power supplies provide industry standard 12, 24 or 48 Vdc. Application requirements often dictate that these supplies withstand harsh environmental conditions.

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DIN RAIL MOUNTING KITS ARE AVAILABLE FOR MANY POWER SUPPLIES. AN EXAMPLE IS THE EB35DIN MOUNTING KIT WHICH CONSISTS OF AN ALUMINUM PLATE WITH TWO DIN CLIPS AND FOUR SCREWS FOR ATTACHING THE PLATE TO THE BOTTOM OF A SUPPLY OR DC-DC CONVERTER. THE KIT LETS THE SUPPLY SNAP ONTO A 35-MM ‘TOP HAT’ TYPE OF DIN RAIL.

ONE ADVANTAGE OF SWITCH-REGULATED SUPPLIES IS THAT THEY CAN FIT IN LOWPROFILE PACKAGES. EXAMPLES INCLUDE THESE 720-W-OUTPUT SUPPLIES HOUSED IN WL7 AND WL9 CASES.

2 • 2021

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POWER ELECTRONICS HANDBOOK LINEAR POWER SUPPLIES ARE TYPICALLY SPECIFIED FOR USES WHERE LOW POWER LINE NOISE AND LOW EMI ARE BOTH PRIORITIES. EXAMPLES INCLUDE THE GOLD BOX INFINITY LINE WHICH PUT OUT UP TO 450 W AND HAVE TYPICAL MAXIMUM RIPPLE COMPONENTS OF 0.75 MV PEAK-TO-PEAK. Many power supply manufacturers provide simple DIN rail mounting kits for many of their power supplies. For instance the EB35DIN mounting kit from Acopian includes an aluminum plate with two DIN clips attached to it. Four screws attach the plate to the bottom of any mini-encapsulated power supply or mini dc-dc converter with screw terminals. The power supply can then be snapped onto a 35mm ‘top hat’ type DIN rail.

LOW NOISE SUPPLIES Electrical devices switching on and off create electrical noise having a spectrum consisting of a fundamental at the switching frequency as well as numerous harmonics. Another source of conducted electromagnetic interference (EMI) is ripple in the converter output. Ripple can be a factor because rectification and filtering of a switching supply’s output causes an ac component that rides the dc output. Typical ripple noise levels are on the order of hundreds of microvolts to tens of millivolts. This level is unacceptable in medical equipment, low-noise amplifiers,

signal processing, data acquisition, automatic test equipment and laboratory test equipment. Linear regulated power supplies have little ripple and little output noise, making them suitable for laboratory, medical and instrumentation applications. Equipment that generates and manages X-ray data, requiring reliable, consistent and clean power is an example. In one case, a linear power supply handled a dark room environment for developing X-ray imagery. All LED indicators and digital meters on the supply could be switched off while the power system was in use; toggle switches on the front panel provided individual control of LED indicators and digital meter backlighting. For the defense industry, components must withstand the extreme environmental conditions that the military faces. In particular, power supply systems must be designed and qualified to strict manufacturing, environmental and operational Military Specifications (MIL-SPEC). MIL-STD-704 (Aircraft Electrical Power Characteristics) for instance, is a U.S. military standard that defines a standard power interface between a military aircraft and its equipment and carriage stores. The standard covers such topics as voltage, frequency, phase, power factor, ripple, maximum current, electrical noise and abnormal conditions for both ac and dc systems. A point to note is the requirements and specifications for defense applications are unique. Such requirements may prevent standard switching power supplies from meeting military design needs. Online tools that power supply manufacturers provide can speed the supply selection process. An example is the online Acopian custom system builder. Engineers can use this tool to easily custom-design dc-dc and ac-dc power supplies with just a few clicks, selecting options that range from the number of outputs to the topology of the supply circuit.

P RoHS

REFERENCES WWW.ACOPIAN.COM ACOPIAN

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A57_DesWrld9x10_875.qxp_Layout 1 12/18/20 9:53 AM Page 1

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POWER ELECTRONICS HANDBOOK

Maximizing the performance of SiC through gate drive techniques NEW GATE DRIVERS SIMPLIFY THE TASK OF DEPLOYING SILICON-CARBIDE FETS TO SHRINK THE SIZE AND BOOST THE EFFICIENCY OF POWER INVERTERS.

The

electric vehicle revolution is upon us. Automotive companies search desperately for a technological edge, investors see EV companies as more valuable than traditional automakers, and governments press ahead with ending sales of internal combustion vehicles. Under the hood, the power electronics driving EVs are rapidly evolving as well. Wide bandgap FET technology like silicon carbide (SiC) promise dramatic efficiency improvements, lighter weight systems, and smaller batteries. In automotive designs, SiC is delivering on these promises and driving innovation in the next generation of EVs. The fundamental advantage of SiC and other widebandgap devices can be understood from their bandgap, the energy difference between the top of the valence band and the bottom of the conduction band . Moving electrons from the low-energy valence band to the higherenergy conduction band causes a material to conduct. To move an electron from the valence band to the conduction band in silicon takes 1.1 eV. SiC, on the other hand, has a bandgap of 3.2 eV, so it takes more energy to move the electrons to the SiC conduction band. This fact translates into a much higher breakdown voltage for a given die size than is the case for silicon devices. The SiC die size advantage and benefits such as lower onresistance (RDSON) and faster switching speed are practically tailor made to solve the biggest EV challenges. Three of the main limitations of EVs are charge time, range, and cost. Boosting the high- voltage portion of the EV inverter circuit, known as the dc link, to 800 or even 1, 000 V lowers the current involved and thus allows lighter-weight cables and magnetics. Higher voltages require switching devices with higher breakdown voltages, often up to 1,200 V. For standard silicon MOSFETs, scaling the breakdown voltage to this level, while maintaining high current capability, is impractical because the necessary die size becomes prohibitive. Bipolar silicon devices, primarily insulated bipolar gate transistors (IGBTs), solve this problem but sacrifice switching speed and limit power conversion efficiency. The wide band gap of SiCs allows unipolar FET devices,

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CHARLIE ICE

SILICON LABS

with dramatically smaller die sizes, to exhibit the same breakdown voltage and current ratings as traditional IGBTs. This ability brings several improvements to power conversion systems while allowing higher dc link voltages and reducing vehicle weight. To improve the range of an EV, either the battery capacity must increase, or the vehicle must become more efficient. In general, boosting the battery capacity adds cost, size, and weight, so designers focus instead on improving the efficiency of the vehicle power conversion systems. With the correct switching devices, designers can up the power supply switching frequency as a way to raise efficiency while simultaneously reducing the size of the magnetic components, reducing cost and vehicle weight. In addition, the resulting highly efficient converters need less heat sinking and cooling systems. SiC FETs naturally accommodate these high switching frequencies because they dissipate little energy in each charge/discharge cycle. Furthermore, the material properties of SiC, combined with the small die sizes, allow operation at higher temperatures with lower losses than IGBTs. Unlike IGBTs, SiC FETs have an RDSON specification, and the rated RDSON varies little with temperature. This concept is critical for high-power EV applications, where the switching devices handle kilowatts of power and routinely reach high temperatures. In addition, IGBTs are typically optimized for maximum current; their conduction losses rise dramatically at less than maximum load. SiC FETs, however,

S i C ON RESISTANCE VS TEMPERATURE

HOW THE R DSON OF THE CREE WOLFSPEED E3M0065090D AUTOMOTIVE SIC FET CHANGES WITH TEMPERATURE.

2 • 2021

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SiC GATE DRIVERS

DRIVER WITH MILLER CLAMP CDG

higher gate drive signal and a negative gate voltage in the off state. The latest isolated gate drivers integrate the features necessary to meet all these requirements. Many high-voltage automotive systems use isolation devices, such as isolated gate drivers, to separate the low-voltage THE INTEGRATED MILLER CLAMP controller from the highON THE SILICON LABS SI828X voltage portion of the ISOLATED GATE DRIVER. system. The high switching frequencies used in most SiC FET designs subject the isolated gate driver to maintain their efficiency at low loads. This fast transients. A gate driver with a commonbehavior is particularly useful in automobiles, mode transient immunity (CMTI) of at least where systems like the traction inverter operate 100 kV/µsec can withstand these transients. at different loads for long periods. Furthermore, the driver’s propagation delay All of these improvements from SiC and channel-to-channel skew generally must FETs add up to greater efficiencies, smaller be below 10 nsec for the design to be stable batteries, lower costs, and ultimately more at such high speeds. As automotive systems capable EVs. Nonetheless, the adoption of SiC push dc link voltages higher, the isolated technology requires designers to learn new gate driver must also have enough maximum techniques, and some of the most important insulation working voltage (VIORM). Thanks to techniques center on the gate driver. technological advances, designers can simply Silicon carbide FETs, with their smaller die select an isolated gate driver that meets the size and higher switching frequencies, require needs of a SiC FET based system. slightly different gate drive techniques. The Many new isolated gate drivers, such as the small die size makes SiC FETs more susceptible Silicon Labs Si828x, also include an integrated to damage, and the higher frequencies Miller clamp and desaturation detection to necessitate a gate driver with improved protect the SiC devices. In a half or full bridge performance. Finally, SiC FETs often require a

GATE DRIVE WITH DESATURATION

THE INTEGRATED DESATURATION CIRCUIT ON THE SILICON LABS SI828X ISOLATED GATE DRIVER.

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configuration, the switching device on the lower half of the bridge experiences a fast change in voltage on the drain when the upper device turns on. This change induces a current in the gate to drain parasitic capacitance which can otherwise discharge through the gate and turn on the lower device. This “Miller parasitic turn on” causes shoot-through which will quickly damage a SiC device. An integrated Miller clamp discharges the gate to drain parasitic capacitance when it reaches a pre-configured threshold. In addition, abnormal load conditions may cause the switching device to fall out of saturation and be damaged. But there is a desaturation circuit integrated into the Silicon Labs Si828x gate driver. If the voltage across the switching device rises above a configured threshold, the gate driver quickly responses and shuts it off gracefully. It does so using a soft shutdown circuit to limit the induced shutdown voltage across the switching device. In the case of a SiC FET, the protection circuit must react quickly, often in under 1.8 µsec, to be effective. With these three features integrated into the gate driver, it becomes much simpler to design a robust, reliable SiC power converter. The final aspect of driving SiC FETs is the use of a negative voltage when turning off the FET. The negative voltage works in conjunction with the Miller clamp to secure the FET in the off state, an aspect of operation that is critical in controlling shoot-through current in high frequency power converters. The methods to generate the necessary negative voltage rail are beyond the scope of this article. However, selection of a gate driver with an integrated dcdc converter generally simplifies the design. All in all, silicon-carbide switches open the door to faster switching speeds, greater efficiency, and higher power density than ever before. Among other benefits, their high breakdown voltage and thermal performance comprise natural building-blocks for automotive power systems. These advantages, coupled with the improved capabilities of isolated gate drivers, make them core technologies in the electrification revolution.

REFERENCES SILICON LABS

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SUPPLY HOLD-UP TIME

How to boost output hold-up time in power supplies MANY SUPPLIES HAVE OUTPUTS THAT DON’T STAY UP LONG ENOUGH IN THE EVENT OF A POWER LOSS. HERE ARE SOME WAYS TO REMEDY THE PROBLEM. TDK-LAMBDA AMERICAS

GENERIC SWITCHING SUPPLY BLOCK DIAGRAM

Rectification & filtering

Switching section

Boost converter

Rectification

Input filtering

AC input

DC output

DC-DC converter

ac-dc power supply’s input voltage can be interrupted during circumstances such as a brown out condition or a brief power failure. When this happens, the dc output will only remain within regulation for a short period of time. The time period is specified on the power supply datasheet as the hold-up time. During this hold-up time the power supply relies on energy stored in its capacitors to maintain operation. In a typical ac-dc supply, the ac input voltage is filtered, rectified and boosted to provide a dc bus voltage of around 390 V. The dc-dc converter section of the supply provides primary-secondary isolation and reduces the 390-V bus down to the eeworldonline.com | designworldonline.com

desired dc output voltage. Typically, there is a high-voltage bus electrolytic capacitor that reduces the ripple voltage on the 390-V bus and stores energy to keep the dc-dc converter operational during brief interruptions to the ac input. Depending on the application, the length of the hold-up time typically varies from a half to a full cycle of the incoming ac 50/60-Hz voltage. This period is typically 8 to 20 msec at 100% load but is normally sufficient to keep the attached electronic equipment from having to restart or reboot. However, some applications demand an extended hold-up time. The medical industry’s concern regarding hold-up time has risen since the release of the EN 606011-2; 2015 (Ed4) immunity standard. Primarily 2 • 2021

A SIMPLE BLOCK DIAGRAM OF A POWER SUPPLY SHOWING DC BUS CAPACITOR C1 AND OUTPUT CAPACITOR C2. created to address the growing number of products used in home healthcare, this standard specifies multiple ac voltage dips ranging from 20 msec to five seconds. The longer outages are generally addressed by including batteries or otherwise ensuring that no harm will befall the patient or operator if the power supply output voltage drops out of the regulation band. Airborne equipment is covered by the DO-160 standard. Section 16 refers to power input, simulating conditions of aircraft DESIGN WORLD — EE NETWORK

POWER ELECTRONICS HANDBOOK EXAMPLE SUPPLY HOLD-UP TIMES 1000

Hold up time (ms)

1000

100

Output current (%) power from before engine start (using auxiliary ground-based power) to after landing, including emergencies. The requirement is for a hold-up time of at least 200 msec.

HOW TO INCREASE HOLD-UP TIME There are various methods to extend power supply hold-up time, each with advantages and disadvantages. The amount of energy stored in a capacitor C = ½ ×C ×V2. To increase that energy storage and hence the hold-up time, either the amount of capacitance or the voltage on the capacitor must rise. Because V is squared, an increase in the value of V will have a greater impact. The following examples are based on a hypothetical system requiring a load of 150 W at 12 V, with a minimum output voltage requirement of 11.5 V as the output decays to zero; 200 msec is the desired hold-up time. 1. Using a higher-rated power supply and operating it at a reduced load. Consider the example of using the 150-W 12-V output TDK-Lambda RWS150B-12 power supply and operating it at 100% load. Based on test data, the supply would exhibit a hold-up of just over 30 msec.

40 60 80 Output current (%)

100

EXAMPLE: HOLD-UP TIME VERSUS OUTPUT LOAD FOR THE RWS150B-12 AND CUS1500M-12 POWER SUPPLIES. As expected, the hold-up time depends on the amount of output load. The greater the load, the quicker the high-voltage bus capacitor’s stored energy depletes. To get a 200-msec hold-up, we could switch to another supply such as the TDK-Lambda 1,500-W- rated CUS1500M-12. This supply has a larger bus capacitance which would provide enough energy to hold-up a 150-W load for over 200 msec. Thus swapping supplies solves the problem, but at the cost of using a much larger, more expensive power supply. 2. Adding capacitance across the power supply output terminals or load. At ﬁrst glance, adding additional capacitance across the output seems like an easy solution. But consider the size of the necessary capacitor. Suppose stored energy E in Joules is the product of the output power Pout in Watts and time t in seconds. Then E = Pout × t. As called out earlier, E is half the product of the capacitance, C, and the voltage V2: E = 0.5×C×V2.

PLACING MORE CAPACITANCE ACROSS THE LOAD ADDING LOAD CAPACITANCE C3 AT THE SUPPLY OUTPUT.

Rectification & filtering

Switching section

Boost converter

Rectification

Input filtering

AC input

DC output

DC-DC converter

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SUPPLY HOLD-UP TIME HIGHER BUS CAPACITANCE TO BOOST HOLD-UP TIME

C1a

Rectification & filtering

Switching section

Boost converter

Rectification

Input filtering

AC input

DC output

DC-DC converter

ADDING HIGH-VOLTAGE BUS CAPACITANCE C1A. Rearranging for C we get C = 2×E/V2. Note: The voltage on capacitor C will drop as it discharges into its load. At some point the voltage will be too low for the load to function, leaving unused energy in the capacitor. That voltage point will be referred to here as Vend. In our scenarios we’ll use a voltage of 11.5 V for Vend. The useful energy for hold-up will be the initial minus the remaining energy: E = (0.5×C×V2) – (0.5×C×Vend2). Factoring the equation results in E = 0.5 × C × (V2 – Vend2) or C = 2×E / (V2 – Vend2). Substituting Pout × t for E we have C = 2 × Pout × t / (V2 – Vend2). Now we can determine the size of the output capacitor needed on a 150-W supply for a 200 msec holdup time. Here, the output capacitor would have to be a massive 4,595,745 uF: C = 2 × Pout × t/ (V2 – Vend2), C = 2 x 150 x 0.18 / (122 – 11.52) = 0.4595745. Note that as the power supply already has 20 msec of holdup capability, t is reduced to 180 msec (0.18 sec).

Even a sufficiently large supercap of this value would consume a large amount of space. One other major concern is the over-current of the power supply during initial turn-on. An uncharged output capacitance would appear to the power supply control circuit as a dead short across the output. The power supply would most likely fail to establish a 12-V output when initially turned on. 3. A customized power supply with a larger high-voltage bus capacitor (C1). As previously mentioned, the energy stored in a capacitor E = 0.5×C×V2, so adding capacitor C1a across the high 390-Vdc bus has a greater effect than adding capacitance across the 12-V output. As the dc bus is not accessible on most power supplies, the adding of capacitors would involve creating a custom or modified standard design. This would involve engineering charges and safety re-certification fees, plus time to make the modification. If the power supply to be modified had a hold-up time of 20 msec, increasing it to 200

msec would require adding the equivalent of nine more C1 capacitors. As C1 typically occupies 5 to 6% of the internal space of a power supply, this strategy would increase the size of the power supply by around 50% for the same product height. A power-module-based approach could also employ a non-isolated ac-dc 360-Vdcoutput converter and a high-voltage input isolated dc-dc 12-V-output converter. This approach again would require a custom board design with engineering and safety certification charges. 4. A 48-V or 60-V-output ac-dc power supply and a wide range input dc-dc converter. For an off-the-shelf standard product, a 48-V-output ac-dc power supply could be used with an isolated wide-range

A SUPPLY PLUS CONVERTER APPROACH: A 48-V OR 60-V POWER SUPPLY AND AN ISOLATED WIDE-RANGE-INPUT DC-DC CONVERTER.

SUPPLY PLUS CONVERTER

AC input

48V or 60V output AC-DC power supply

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Isolated DC-DC converter (wide range input)

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12VDC output

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SUPPLY HOLD-UP TIME ALTERNATIVE SUPPLY PLUS CONVERTER CONFIGURATION

AC input

48V output AC-DC power supply

ANOTHER SUPPLY PLUS CONVERTER APPROACH: A 48-V POWER SUPPLY AND A NON-ISOLATED DC-DC CONVERTER. input 18-to-75-V dc-dc converter with a 12-V output. The dc-dc converter may require a heatsink or cold plate for cooling. In addition to providing the hold-up inside the 48-V output ac-dc supply, capacitor C1 can be used for additional energy storage. We have taken advantage of the energy storage formula 0.5×C×V2 as V is now 48 V rather than 12 V. The efficiency of the dc-dc converter is considered to be 90%, typical for high-quality converters. The hold-up of the dc-dc supply is 20 msec, requiring an additional 180 msec (t): C = 2 ×P × t / (Eff x (V2 – Vend2)). For our example, the required value of C1 would be 2 × 150 × 0.18 / (0.9 × (482 – 182)) = 30,300 uF.

Non-isolated DC-DC converter (wide range input)

If the 48-V ac-dc supply was replaced by one having a 60-V output, the additional capacitance could be reduced to 18,315 uF. A 72-V output power supply would further reduce that to 12,345 uF. 5. A 48-V output ac-dc supply and a widerange-input non-isolated dc-dc converter. In this case, the chosen dc-dc converter is nonisolated (for higher efficiency, smaller size and lower cost) and has a wide input range of 9 to 53 V. A non-isolated dc-dc converter would not require heatsinking. The efficiency of the non-isolated dc-dc converter is assumed to be 96%. The hold-up of the ac-dc supply is 20 msec, requiring an additional 180 msec (t): C = 2 × P × t / (Eff x (V2 – Vend2)). For our example, the required value of C1 would be 2 ×150 × 0.18 / (0.96 × (482 – 92)) = 25,304 uF.

12VDC output

REFERENCES TDK-LAMBDA AMERICAS WWW.US.LAMBDA.TDK.COM

METHOD

ADDITIONAL CAPACITANCE PROS

CONS

Larger power supply

None

Standard part

Cost and size of bigger supply

Larger output cap

4,595,745uF

Standard part

Size and potential start-up issues

Larger HV bus cap

9 x C1 (~9,000uF)

Custom solution

Eng. cost and development time

48V AC-DC + isolated DC-DC

30,300uF

Standard parts

Space for DC-DC and heatsinking

60V AC-DC + isolated DC-DC

18,315uF

Uncommon 60V AC-DC Standard DC-DC

Space for DC-DC and heatsinking

72V AC-DC + isolated DC-DC

12,345uF

Uncommon 72V AC-DC Standard DC-DC

Space for DC-DC and heatsinking

48V AC-DC + non-isolated DC-DC

25,345uF

Standard parts

Space for DC-DC

THE PROS AND CONS OF TYPICAL APPROACHES AIMED AT BOOSTING SUPPLY HOLD-UP TIME.

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POWER ELECTRONICS HANDBOOK

Super-efficient power converters help hovering drones take flight

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POWERING DRONES

When

the task calls for a aerial system that can hover in place, the often-chosen approach is to employ a tethered drone. A tethered drone, generally a multirotor, gets its power and communications signals via a ﬂexible wire or cable. Where battery powered multirotor drones can exhaust their batteries in 20 minutes or so, tethered drones receive their power through the tether from a base station. This enables them to stay aloft for hours or days.

VISIBLE ON THE UMAR DRONE ARE THE EIGHT ROTOR MOTORS AND THE TETHER THAT SUPPLIES POWER AND COMMUNICATIONS TO THE UAV.

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One limiting factor of the scheme is the weight of the tether. Every ounce of tether weight is one less ounce available for the drone payload. So drone tethers may be made of aramid or other lightweight synthetic materials to provide strength, with copper or plated copper for power conduction and optical fiber for data and communications. For similar reasons, designers endeavor to keep the tether power wires light weight. So the usual approach is to use a high voltage/low current supply at the base station to minimize the size of the tether conductors, then down-convert the voltage on the UAV. This is the approach used by DPI UAV Systems in Essington, Pa. on its Unmanned Multirotor Aerial Relay (UMAR). UMAR is a tethered drone that lifts 15-lb payloads, such as communication antennas, up to 500+ ft in the air. By levitating an antenna, the UMAR system can extend boat/ship radio line-of-site from eight miles to 30 miles, thereby boosting communication range. UMAR consumes 8 to 10 kW, all of which is supplied via the tether. DPI down-converts 800-V power coming up through the tether into 50 V via Vicor Ultra High-Voltage (UHV) BCM VIA modules. Eight BCMs power the UMAR’s eight independent rotors, with the ability to share power among the rotors in parallel for redundancy. An additional onboard UHV BCM powers the avionics, autopilot and payload. The UHV BCM4414 modules are fixed-ratio dc-dc converters providing high-efficiency conversion (98%) — higher efficiency than would be available from regulated converters. Measuring just 4.35x1.40x0.37 in (110.55x35.54x9.40 mm), these modules obey PM Bus commands and provide up to 776 W/in3 power density while incorporating integrated EMI filtering. The low-profile, flat-sided power modules sit in two waterproof--to prevent moisture ingress from saltwater and rain--aluminum enclosures and mount to the host PCBs in an architecture that allows two-sided cooling. A system of heat pipes and heat sinks provide passive cooling when the drone is on the host vessel. When airborne, the spinning rotors provide additional active cooling. 2 • 2021

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POWER ELECTRONICS HANDBOOK

A low-voltage-side-referenced PM Bus-compatible telemetry and control interface provides access to fault monitoring and other telemetry functions. This PM Bus capability allows UMAR operators to monitor system temperatures, voltage and currents in real time, information which is particularly valuable in hot climates. Integrated EMI filtering built into the Vicor UHV BCMs helps minimize electrical noise that might otherwise garble RF communications between the drone and host or support vessels. Minimization of conducted EMI on the tether is especially important when the drone operates near radio equipment. UHV BCMs provide a clean EMI signature with few harmonics. A small companion filter reference design outside the VIA package cuts conducted EMI levels to well below typical requirements. “Using Vicor power modules, we have been able to lower the weight of all components onboard the drone to increase altitude and airspeed while carrying the required mission payload,” said Joe Pawelczyk, Vice President of Operations at DPI. “Nobody else really has the power density of Vicor, so we can achieve top levels of maneuverability, performance and hover control with their components. This enables DPI tethered drones to lift more, fly higher, and fly faster.“ UMAR is designed to maintain stationary positioning in severe maritime storm conditions. The task is often complicated by turbulent waters that toss around the host vessel. So the drone requires the power capacity and agility to accelerate rotor speeds for lift and yaw in short or prolonged bursts to maintain altitude – with instantaneous responsiveness.

THE POWER SYSTEM FOR THE UMAR: THE BASE STATION GENERATES POWER AT 800 V WHICH GETS FED UP THE TETHER TO THE DRONE. THERE, EIGHT VICOR FIXED-RATIO CONVERTERS EACH FEED 48 V/130 A TO ONE OF THE ROTOR MOTORS. (NOT ILLUSTRATED: THE VICOR CONVERTERS ARE WIRED SO THEY CAN SHARE POWER IF NEED BE.) A NINTH VICOR CONVERTER GENERATES 48 V/10 A TO RUN THE UMAR AVIONICS.

Tested extensively in real-world operating conditions, DPI tethered drones are currently being qualified for use by the U.S. Navy in marine and maritime environments for intelligence, surveillance, reconnaissance (ISR), communications and video applications. The system has been designed to withstand harsh temperatures A UMAR IN USE AT SEA. THE from zero to 120°F. These drones can provide 400+ hours of nonstop up time DRONE IS SMART ENOUGH TO and operations. AUTOMATICALLY TRACK THE BASE In addition to its current trial STATION AS IT MOVES AROUND AND deployments with the U.S. Navy, DPI IS BUFFETED BY WAVES. technology is being evaluated by government agencies, contractors and other entities. It has shown significant promise for additional applications like first-response disaster relief and largearea monitoring (public events, stadium security, etc.). Anywhere a hover-in-place communications and surveillance presence might be needed, DPI multirotor drones could be readily deployed.

REFERENCES VICOR CORP.

WWW.VICORPOWER.COM

DPI UAV SYSTEMS WWW.DRAGONFLYPICTURES.COM/

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POWER ELECTRONICS HANDBOOK

The changing maze of power supply standards BREXIT ALTERED THE LANDSCAPE OF STANDARDS FOR POWER SUPPLY PRODUCTS. HERE’S AN UPDATE ON THE LATEST REVISIONS.

RON STULL

CUI INC.

COMMON IEC SAFETY STANDARDS

IEC 60601-1 Medical and dental equipment

IEC 62368-1

IEC 60335-1

Information communication technology and audio visual products

Household electrical appliances

The

usual way of choosing a power supply is to search on the electrical performance parameters the application demands. But today, it is actually more efficient to start with the national or international standards the end product must meet. Standards requirements can vary markedly among domestic, IT, audio, industrial and medical applications. And it can’t be assumed that compliance with what might be considered a ‘high-end’ application automatically is acceptable for more commercial end-uses. For example, a medically certified supply requires input fusing in both line and neutral for Class I applications (with a protective ground). But in domestic equipment, this kind of fusing is expressly forbidden.

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IEC 60335-1

Transformers, reactors, and power supply units

Equipment manufacturers are also increasingly required to assess the risks of their products according to ‘hazard-based’ standards, which include an evaluation of where the product might be used. For example, a commercial laptop that is sold as ruggedized with an add-on rubber casing might reasonably be expected to be used in industrial or even outdoor environments, requiring its external charger be certified to a more specific standard. Aspects controlled by standards fall into three main areas: safety, electromagnetic compatibility (EMC) and energy efficiency. Common safety standards that apply are set by the International Electrotechnical Commission (IEC). Other standards apply for particular applications such as hazardous locations, railroad, test equipment, and so forth. Some areas haven’t yet standardized. For example, automotive on-board electrical safety currently relies on OEM internal rules and recommendations

2 • 2021

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POWER SUPPLY STANDARDS ALTITUDE ADJUSTMENTS FOR POWER SUPPLIES (FROM IEC 62368-1) Altitude (m)

Normal barometric pressure (kPa)

Multiplication factor for clearances

2000

3000

Multiplication factor for electric strength test voltages < 1mm

1mm to < 10mm

10mm to < 100mm

1.00

1.14

1.05

1.07

1.10

4000

1.29

1.10

1.15

1.20

5000

1.48

1.16

1.24

1.33

Linear interpolation may be used between the nearest two points, the calculated minimum multiplication factor being rounded up to the next higher 0.01 increment.

from automotive industry associations such as the Society of Automotive Engineers (SAE). In addition, IEC standards are only a basis. Different countries or economic areas may choose to accept them, or not, and sometimes with local variations. One example: IEC 62368-1 becoming UL 62368-1 or EN 62368-1. Some countries require compliance with older versions of standards which may even have been withdrawn by the IEC. For example, IEC 62368-1 supersedes IEC 60950-1 which was withdrawn on Dec. 20, 2020. But countries such as China, Korea and Taiwan have yet to publish their versions of IEC 62368-1. They may require that imported products continue to comply and be marked to 60950-1 or comply with their standards, such as GB 4943.12011 and GB 8898-2011 in China. IEC 62368-1 is already at its third edition with the fourth a work in progress. Countries outside the European Union may require compliance with the third edition. But within the EU, this edition is not formally published in the ‘Official Journal’ and therefore cannot be used as a basis for CE marking. The fourth edition is unlikely to be published until late 2021 or 2022. But as its contents become known, power supply manufacturers are likely to pre-empt the standard and ‘design to meet’ before it becomes mandatory. To complicate matters further, different countries have different ‘grandfathering’ schemes allowing the continued sale of products after the withdrawal date of older standards. This is specifically allowed in the second edition of IEC 62368 (clause 4.1.1) for power supplies as components within end applications that meet IEC 60950-1. The third edition, however, removes this concession.

SPECIAL POWER SUPPLY CONSTRUCTION Geography can affect standards requirements as well. Later standards use altitude as an environmental limit that affects requirements for clearances and isolation test voltages. In IEC 62368-1 for example, a maximum of 2,000 m is assumed with safety clearance multiplied by a factor of 1.48 for operation up to 5,000 m. This requirement is not exceptional; eight world capitals including Mexico City sit at an elevation over 2,000-m. Smaller eeworldonline.com | designworldonline.com

towns and cities including STANDARDS SPECIFY THAT La Rinconada in Peru POWER SUPPLY DESIGNS MUST (population 30,000) sit at over 5,000 m. Power ALLOW FOR ALTITUDE. SOURCE: supplies manufactured or ADAPTED FROM IEC 62368-1. sold in China according to GB 4943.1-2011 for domestic use must all be rated for 5,000-m operation unless specifically marked otherwise. Products may also require thermal derated at high altitude to maintain their safety certification, due to the reduced heat transfer capacity of thin air. Standards such as IEC 62368-1 also contain other conditions of use; ac supplies are defined to have ‘overvoltage’ (OV) categories and environments have ‘pollution degrees’ (PD). Thus a claim of ‘IEC 62368-1 certification’ should state the actual OV category and PD achieved in the design. Compared with the plethora of safety standards, there are only a few standards for electromagnetic compatibility (EMC), and they have been consolidated in recent years. In Europe, for example, EMC standards for emissions from IT equipment and both professional and domestic audio-visual equipment were combined in 2017 into one standard, EN 55032. This standard calls up the limits and test methodologies defined in CISPR 32 (CISPR stands for the French words for the Special Committee on Radio Interference. It publishes a number of EMC standards used for a variety of product families}. Top-level IEC standards for EMC are in the IEC 61000 series covering conducted and radiated emissions, immunity to RF and magnetic fields, surges, transients, dips, induced line harmonic currents, and electrostatic discharge (ESD). As with safety, IEC publications don’t cover automotive EMC standards but documents from the International Standards Organization (ISO) in the ISO 114542 series cover test methods for immunity testing. Again, the SAE publishes a range of documents covering the same areas. EMC standards incorporate different limit levels depending on the environment or application. Many engineers are familiar with the ‘Class A’ and ‘Class B’ emissions limits for industry and domestic use, but there is more to consider. For example, in the EN 61000 series, ‘severity’ levels 2 • 2021

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POWER ELECTRONICS HANDBOOK SINGLE-VOLTAGE EXTERNAL AC-DC POWER SUPPLY, BASIC-VOLTAGE A SUMMARY OF DOE LEVEL VI EFFICIENCY AND NO-LOAD LOSS LIMITS. are defined for immunity to ESD and conducted and radiated interference. Effects ‘criteria’ are also specified, ranging from ‘A’ for ‘no effect’ to ‘D’ for permanent loss of function or damage. The acceptable level depends on the application; critical medical equipment might demand uninterrupted operation with the highest severity level of interference. But a disposable cell phone adapter might be fine with a non-resettable fuse opening inside after a mains surge. In both scenarios, the products could be said to be EMC compliant but with very different outcomes. Medical power supplies generally, whether in ‘operator’ or ‘patient connect’ applications, have enhanced EMC compliance requirements in the fourth edition of IEC 60601-1- 2, based on the IEC 61000 series but with higher severity levels or expanded scope.

POWER CONVERTER EFFICIENCY The efficiency of power converters at full and light load, along with no-load losses, has long been an economic concern. Environmental impact worries are now mandating tighter efficiency limits. Standards defined by the U.S. Dept. of Energy are at Level VI. In the E.U., the Ecodesign Directive 2019/1782 sets mandatory limits substantially the

Nameplate output power (Pout)

Nameplate output power (Pout) minimum efﬁciency in active mode (expressed as a decimal)

Maximum power in no-load mode (w)

Pout ≤ 1 W

≥ 0.5 x Pout + 0.16

≤ 0.100

1 W < Pout ≤ 49 W

≥ 0.071 x In(Pout) - 0.0014 x Pout + 0.67

≤ 0.100

49 W < Pout ≤ 250 W

≥ 0.880

≤ 0.210

Pout > 250 W

≥ 0.875

≤ 0.500

SINGLE-VOLTAGE EXTERNAL AC-DC POWER SUPPLY, LOW-VOLTAGE Nameplate output power (Pout)

Nameplate output power (Pout) minimum efﬁciency in active mode (expressed as a decimal)

Maximum power in no-load mode (w)

Pout ≤ 1 W

≥ 0.517 x Pout + 0.087

≤ 0.100

1 W < Pout ≤ 49 W

≥ 0.0834 x In(Pout) - 0.0014 x Pout + 0.609

≤ 0.100

49 W < Pout ≤ 250 W

≥ 0.870

≤ 0.210

Pout > 250 W

≥ 0.875

≤ 0.500

SINGLE-VOLTAGE EXTERNAL AC-AC POWER SUPPLY, BASIC-VOLTAGE Nameplate output power (Pout)

Nameplate output power (Pout) minimum efﬁciency in active mode (expressed as a decimal)

Maximum power in no-load mode (w)

Pout ≤ 1 W

≥ 0.5 x Pout + 0.16

≤ 0.210

1 W < Pout ≤ 49 W

≥ 0.071 x In(Pout) - 0.0014 x Pout + 0.67

≤ 0.210

49 W < Pout ≤ 250 W

≥ 0.880

≤ 0.210

Pout > 250 W

≥ 0.875

≤ 0.500

SINGLE-VOLTAGE EXTERNAL AC-AC POWER SUPPLY, LOW-VOLTAGE Nameplate output power (Pout)

Nameplate output power (Pout) minimum efﬁciency in active mode (expressed as a decimal)

Maximum power in no-load mode (w)

Pout ≤ 1 W

≥ 0.517 x Pout + 0.087

≤ 0.210

1 W < Pout ≤ 49 W

≥ 0.0834 x In(Pout) - 0.0014 x Pout + 0.609

≤ 0.210

49 W < Pout ≤ 250 W

≥ 0.870

≤ 0.210

Pout > 250 W

≥ 0.875

≤ 0.500

MULTIPLE-VOLTAGE EXTERNAL POWER SUPPLY

THE NEW UKCA MARK. 34

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Nameplate output power (Pout)

Nameplate output power (Pout) minimum efﬁciency in active mode (expressed as a decimal)

Maximum power in no-load mode (w)

Pout ≤ 1 W

≥ 0.497 x Pout + 0.067

≤ 0.300

1 W < Pout ≤ 49 W

≥ 0.075 x In(Pout) + 0.561

≤ 0.300

49 W < Pout ≤ 250 W

≥ 0.860

≤ 0.300

2 • 2021

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POWER SUPPLY STANDARDS

same as those of the DoE with some extra labelling and documentation requirements. Outside the U.S. and E.U., countries mostly use the DoE speciﬁcations, sometimes mandatory at lower levels with voluntary compliance at higher levels. The CE mark is an indication that a product meets all requirements for unrestricted sale throughout Europe. These requirements are set by European Directives and relate to safety, health and environmental impact. Marking is not required for all product categories, and in many cases, manufacturers can self-declare they comply. For common power supply products, the Low Voltage and EMC directives are most relevant referring to Euronorms such as EN 62368-1 for safety. But other standards may apply depending on the end-use, such as EN 60601-1 for medical applications. Of course, certiﬁcation and marking of power supplies changed in Jan. 2021 for manufacturers that sell into the UK and Northern Ireland. With the UK withdrawal from the European Union, a new UKCA mark

is required to certify the product meets the relevant standards, similar to the CE mark. Initially, the same international standards will be used, with a BS preﬁx. But there is no guarantee the standards will remain in lock-step with IEC versions in future revisions. Northern Ireland now requires dual marking, CE and UK(NI), although longer-term requirements are unclear, subject to UK-EU trade negotiations. Fortunately, many power supply manufacturers have products holding worldwide certiﬁcations for just about any application. Working with such suppliers can take the stress out of choosing a power supply.

REFERENCES WWW.CUI.COM CUI INC

Proven integrity AND industry know-how Electrocube is one of the most respected design manufacturers of passive electrical component products for a wide range of standard and custom applications – from aerospace and audio to elevators and heavy equipment – as a capacitor supplier, resistor-capacitor distributor, and more.

Bishop Electronics, Seacor, Southern Electronics, F-Dyne

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POWER ELECTRONICS HANDBOOK

Selecting dc-link Capacitors for inverters ONE KEY FACTOR: DETERMINING THE NUANCES OF HOW CAPACITORS HANDLE EXPECTED RIPPLE CURRENTS.

Examine

a dc link capacitor’s ac ripple current and you’ll realize it arises from two main contributors: the incoming current from the energy source and the current drawn by the inverter. Of course, capacitors cannot pass dc current; thus, dc current only flows from the source to the inverter, bypassing the capacitor. Power factor correction (PFC) in the converter and/or regenerative energy flow in certain inverter topologies can complicate matters. But in all cases, instantaneous current is conserved at the three-current node of the dc link capacitor connection. Although some cancellation can arise between the ac components of the source current and the inverter current, it is usually a good approximation or at least conservative to estimate the capacitor’s RMS ripple current as I2capRMS ~ I2sourceRMS + I2inverterRMS

(1)

This is usually valid because the converter stage generally has much lower frequency ripple current content than the inverter stage. The approach to the analysis we’ll use starts by examining the converter stage alone and treating the inverter as a load with a fixed power dissipation or resistance. To simplify things and generalize the conclusions, we’ll implement a Per-Unit (PU) analysis. (As a quick review, a perunit system expresses quantities as fractions of a defined base unit quantity. This simplifies calculations because quantities expressed as per-unit do not change when they are referred from one side of a transformer to the other.) We’ll use as a base the load power drawn by the inverter, assuming a conserved quantity, power, P; and a mains frequency, f. This is basically the ideal dc power delivered to a load resistor, so the base voltage is equal to the peak voltage at zero ripple voltage. The relationships are: Vbase = Vdc(Peak), Ibase = P/Vbase, Rbase = V2base/P, Lbase = V2base/2πfP, Cbase = P/2πfV2base Now consider a rectified single-phase 50-Hz mains with ideal diodes. Such “linear” power supply schemes can produce a high ripple current in the dc link capacitor which here serves as a filter capacitor. The current pulses charging the capacitor when the diodes

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2 • 2021

SAM G. PARLER, JR., P.E.

CORNELL DUBILIER

POWER FLOW IN A TYPICAL INVERTER

Converter

Inverter DC link converter

IN GENERAL, THE POWER FLOW OF A VOLTAGE-SOURCE INVERTER (VSI) IS LEFT-TO-RIGHT UNLESS POWER FACTOR CORRECTION (PFC) OR REGENERATIVE SCHEMES ARE PRESENT.

FULL-WAVE BRIDGE A TYPICAL FULL-WAVE BRIDGE SCHEMATIC WITH LINE INDUCTOR, FILTER CAPACITOR, AND RESISTIVE LOAD.

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LINK CAPACITORS SINGLE-PHASE FULL-WAVE BRIDGE PER-UNIT RIPPLE VOLTAGE

SINGLE-PHASE FULL-WAVE BRIDGE PER-UNIT I cap,RMS

are forward-biased are generally much shorter than the time over which the capacitor discharges into the load. From the principle of charge conservation in a capacitor, these pulses are therefore quite a bit higher in amplitude than the load current. The high pulses usually result in the capacitor RMS ripple current exceeding the dc current delivered to the load. Most power supply designers want a peak-to-peak ripple voltage below 5% and usually limit line inductance to about 5% per-unit. A Spice analysis reveals that a single-phase fullwave bridge requires a lot of capacitance, on the order of 40 PU or more. As long as some line inductance (such as 1% per-unit) is incorporated, the RMS ripple current is relatively insensitive to the level of capacitance. Plots of capacitor ripple current reveal only few frequency components at two, four, and six times the line frequency. Now consider a half-wave bridge. It is even more demanding on a per-unit basis than a full-wave bridge, with regard to the capacitor ac RMS ripple current and peak-to-peak ripple voltage. It takes a capacitance on the order of 100 PU or more to realize a peak-to-peak ripple voltage of less than 5%-It’s probably cheaper to just add three diodes! Here, it takes more line inductance--several percent per unit--to lower the RMS ripple current to a modest level. The frequency content of the capacitor ripple current is nearly zero at dc (0 Hz) as it must be. There are only a few frequency components at half the frequencies of the full-wave bridge at one, two, and three times the line frequency, rolling off rapidly. Now consider three-phase, six-diode rectifiers. The per-unit inductance is in each leg of the three-phase lines. We keep the same base units as for single-phase so the comparisons will be on the nominal power delivered to the resistive load at its nominal peak voltage. Such a rectified waveform, without L and C, would

HALF-WAVE BRIDGE

PER-UNIT ANALYSIS EXAMPLES FOR A FULL-WAVE BRIDGE. TOP, PERCENT PEAK-TO-PEAK RIPPLE VOLTAGE VERSUS LINE INDUCTANCE FOR FOUR VALUES OF FILTER CAPACITANCE. BELOW, RMS RIPPLE CURRENT THROUGH THE FILTER CAPACITOR VERSUS LINE INDUCTANCE FOR FOUR VALUES OF FILTER CAPACITANCE.

A TYPICAL HALF-WAVE BRIDGE WITH LINE INDUCTOR, FILTER CAPACITOR, AND RESISTIVE LOAD.

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POWER ELECTRONICS HANDBOOK SINGLE-PHASE HALF-WAVE BRIDGE PER-UNIT RIPPLE VOLTAGE

SINGLE-PHASE HALF-WAVE BRIDGE PER-UNIT I cap, RMS

PER-UNIT ANALYSIS EXAMPLES FOR A HALF-WAVE BRIDGE. TOP, PERCENT PEAK-TO-PEAK RIPPLE VOLTAGE VERSUS LINE INDUCTANCE FOR THREE VALUES OF FILTER CAPACITANCE. BELOW, RMS RIPPLE CURRENT THROUGH THE FILTER CAPACITOR VERSUS LINE INDUCTANCE FOR THREE VALUES OF FILTER CAPACITANCE.

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2 • 2021

have a ripple voltage below 1-√3/2 ≈ 0.134 per-unit, as this is the maximum droop from the peaks of the three 120º phase-shifted, overlapping sinusoidal mains-voltage rectified waveforms. We observe that with only 1.5% line inductance, ripple voltage can be less than 5% with a capacitance of only four per-unit, as compared to 40 for a single-phase full-wave bridge. However, even for the three-phase, sixdiode rectifier, a Cpu below four isn’t advisable for normal values of Lpu because enormous ripple voltage occurs at 1 and 2 PU. This is due to the LC ringing. Capacitor ESR drops with rising frequency. Instead of ripple current components at small multiples of the utility mains frequency, the peaks on the spectral plot are now at 6, 12, and 18 times the mains frequency. The peaks are at multiples of 25 Hz, and thus the energy bands are at all integer multiples of six times the 50 Hz mains frequency, decaying rapidly. The rectified-mains analysis reveals the energy source to be a sequence of current pulses applied to the capacitor. This suggests that a similar approach may be applied to the analysis of the inverter input current, which draws a sequence of pulses from the dc link capacitor. Both of these sets of pulses will cause voltage ripple as well as ripple current and its attendant heating. As far as the effect on capacitor ripple current and ripple voltage, the main difference between these two distinct sets of pulses, energy source versus inverter sink, is the range of frequencies involved. Typically, the rectified mains and its harmonics are less than 2 kHz, while the inverter switching frequency and its harmonics are usually above 2 kHz. No rule states that the energy source must be diode-rectified. In fact, it could be chopped with IGBTs or SiC switches. Circuits that accomplish power factor correction, bi-directional energy flow (e.g. regenerative braking), etc. generally operate this way. Ultimately the overall analysis of the capacitor ripple current and voltage will involve the superposition of the current flows at its connection node. In general, the inverter stage uses solid-state switches to chop its dc voltage input to create a digital-looking (multilevel) or an even simpler binary (two-level) output voltage waveform, depending upon how many “levels” (discrete voltage values, varying from two to six or more levels) the PWM topology incorporates. (For this reason, these circuits were once referred to as choppers and sometimes still are, especially for dc-dc converters. More formally they are called PWMs, pulse-width modulators.) The unfiltered PWM output voltage is never a true sine wave. But when driving an inductive load such as a motor, the current will tend to be proportional to the time integral of the PWM voltage waveform, whose modulation scheme is designed so the Webers (volt-seconds) of these pulses will produce approximately a sinusoidal current eeworldonline.com | designworldonline.com

LINK CAPACITORS (since, for an inductor L, IL= ∫ v dt/L). And for applications such as a UPS requiring something close to a sinusoidal voltage, an LC or ferro-resonant filter after the PWM stage can effectively integrate and low-pass-filter the voltage waveform. The PWM control/modulation scheme can affect capacitor heating. But usually, power supply designers primarily want to meet goals involving efficiency, cost, size, reliability, total harmonic distortion limits, and sometimes input power factor limits. So minimization of capacitor losses isn’t always the highest priority. Still, it is good to investigate and quantify the relative impact of various factors affecting the capacitor stress. There are many inverter PWM switching and control schemes. Some are carrier-based, some are not. We will consider a somewhat simplified scheme to demonstrate how a typical inverter input affects the dc-link capacitor ripple current and ripple voltage. The scheme we will consider is carrier-based sinusoidal PWM, also known as SPWM. Here, the sinusoidal ac voltage is compared with a high-frequency triangular carrier wave in real time to determine switching states for each pole in the inverter, thereby generating a binary PWM signal. The triangle wave is a fixed-frequency carrier with a repetition frequency equal to the inverter switching frequency. The amplitude of the sinusoidal reference signal as a factor of the amplitude of the reference signal is known as the modulation index, m. Now again consider the three-phase, six-diode rectified 50 Hz mains. Instead of a fixed-resistance load, suppose there is an inductive load with a series resistor. Further suppose we drive this load with a mathematically generated SPWM source voltage. The spectral plot of the inverter input current shows a large dc (0 Hz) component as well as 40, 80, and slight 120 Hz peaks, which are multiples of the 40 Hz single-phase motor drive output frequency. However, there is no sign of the 300-Hz rectified mains current component. On the other hand, the capacitor ripple current shows no dc component but possesses a 40-Hz output current and its multiples, along with a large 300Hz component due to the rectified three-phase 50 Hz mains. Also, the capacitor’s ripple current spectrum contains two sidebands straddling the 300 Hz component; these are at 300 ± 40 Hz = 260 and 340 Hz, typical of the modulated interaction between the mains input and the motor drive output. Note that the existence of such modulated sidebands suggests the possibility that inverter schemes with multiple switching, rectification, and ac-output frequencies can potentially produce ripple current components at frequencies below any of the fundamental frequencies associated with these components, thereby potentially increasing capacitor losses. Capacitor ESR can be modeled approximately as having two terms, a first-term Ro that doesn’t vary with frequency, and a second term Rd which arises from the dielectric loss angle. The series resistance associated with dielectric loss varies approximately in direct proportion to 1/f over an extremely broad frequency range: ESR=Ro+Rd (f) = Ro+DXc = Ro+D/2πfC eeworldonline.com | designworldonline.com

THREE-PHASE SIX-DIODE BRIDGE PER-UNIT RIPPLE VOLTAGE

THREE-PHASE SIX-DIODE BRIDGE PER-UNIT I cap,RMS

PER-UNIT ANALYSIS EXAMPLES FOR THE SIX-DIODE BRIDGE. TOP, PERCENT PEAKTO-PEAK RIPPLE VOLTAGE VERSUS LINE INDUCTANCE FOR FIVE VALUES OF FILTER CAPACITANCE. THERE IS RESONANCE NEAR L PUC PU≈0.01. BELOW, RMS RIPPLE CURRENT THROUGH THE FILTER CAPACITOR VERSUS LINE INDUCTANCE FOR FIVE VALUES OF FILTER CAPACITANCE. CAPACITOR RIPPLE CURRENT PER-UNIT IS LESS THAN HALF THAT OF THE SINGLE-PHASE FULL-WAVE BRIDGE RECTIFIER.

(2) 2 • 2021

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POWER ELECTRONICS HANDBOOK THREE-PHASE BRIDGE

THREE-PHASE, SIX-DIODE RECTIFIED BRIDGE WITH LINE INDUCTOR, FILTER CAPACITOR, AND RESISTIVE LOAD. the contiguous pulse pair. This shows that the dielectric loss component of the dc link capacitor is a strong function of the inverter current pulse spacing. For the case of a PWM inverter with a balanced three-phase output, there is an expression that gives a good estimate of the capacitor ripple current in terms of both the previously discussed modulation index m and the load’s phase current and phase angle φ. It is generally accurate within several percent for most PWM inverter modulation schemes with three-phase ac outputs:

Icap, rms

The second term is equal to the dissipation factor D multiplied by the capacitive reactance. Thus, ESR overall tends to drop monotonically with rising frequency. The dielectric resistance is directly proportional to D and inversely proportional to the product of frequency and capacitance. Thus for a given RMS component magnitude, the lowfrequency components of ripple current cause more heating than the high-frequency components.

SPECTRAL ANALYSIS AND JOULE HEATING Consider the following hypothesis about the current drawn by the inverter from the capacitor: For a given ac RMS value and pulse duty, the dielectric loss component of the dc link capacitor is a strong function of the inverter switching frequency and current pulse spacing, but not of the exact pulse shape. To evaluate the hypothesis, we will examine five pulse-current waveforms, each at a repetition frequency of 1 kHz (i.e. a 1-msec period) and having a value of 33.3 A RMS. First, consider a 100-A, 100-µsec pulse. It has a large fundamental component at the 1 kHz rep rate. Because capacitor ESR has a 1/f term, the dielectric loss would be half as much at 2 kHz as at 1 kHz. This shows that the dielectric loss is a strong function of the switching frequency. With a 130-µF capacitor, a dielectric loss tangent of D = 2% yields a power loss of 11.9 W for this waveform. Next, suppose the RMS value, repetition frequency, and pulse width remain the same, but the pulse shape changes from flat-top to sloped and then to sawtooth--from completely flat-topped to progressively more sloped. Different as the waveforms appear in the time domain, their spectra are similar, and the reference capacitor losses for the three cases are, respectively, 11.9, 11.7, and 10.1 W. This shows that the dielectric loss is not a strong function of the exact pulse shape for a given RMS value and pulse duty. Finally, consider the spectrum and dielectric heating from a 100-µsec-wide flat-top pulse with an inverted twin pulse in two configurations: first, in close proximity and then separated by some distance. The contiguous-pulse arrangement depresses the 1-kHz fundamental harmonic which results in a dielectric heating of the reference capacitor of only 7.46 W. Equal/maximum inter-pulse spacing within the 1-msec period results only in odd harmonics with a pronounced, dielectric-loss-inducing fundamental. At 14.43 W, the power loss in the reference capacitor is nearly twice that of

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IL,rms √2m [√3/4π + (√3/π – 9m/16) cos2 (φ)]

(3)

Note the expression is the ratio of the capacitor RMS current to the line output current from the inverter to the load. Note also that it is independent of the inverter switching frequency. It appears that, for most inverter applications, the ripple voltage can be estimated using a per-unit analysis to pick a range of possible capacitances versus the design’s operating voltage, power level, and frequency. However, are any ride-through requirements may force the use of a higher capacitance level. The capacitor voltage rating must exceed the worst-case peak bus voltage as might arise under “high-line” mains conditions, maximum solar-panel output voltage, etc. Low-ESR aluminum electrolytic capacitors are rated only up to 500 Vdc, so they may need to be connected in series with balancing resistors. Film capacitors are rated to much higher voltages than aluminum electrolytic capacitors and generally do not require a series connection. Aluminum electrolytic capacitors are less expensive per unit of nameplate energy, but they don’t handle as much ripple current per unit of stored energy, so the ripple current handling needs to be investigated. Now consider three-phase inverters at any dc bus voltage. For films and electrolytics, respectively, a rule of thumb is that about five (film) and 50 (electrolytic) mC of capacitor nameplate CV rating (i.e. its volumetric efficiency) is necessary per Amp of ripple current. For example, on a 10-hp motor drive with a 700-Vdc bus, a capacitor ripple current of 7 A RMS

GENERAL VOLTAGE SOURCE INVERTER

GENERAL BLOCK DIAGRAM OF A VOLTAGE SOURCE INVERTER. THE FIRST THREE BLOCKS MAKE UP THE CONVERTER AND THE DC LINK. THE REST CONSTITUTE THE INVERTER STAGE.

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LINK CAPACITORS ELECTRIC VEHICLE MOTOR DRIVE INVERTER

ABS601 & MBS601 600 W AC-DC Convection Cooled Sealed Power Supplies for Harsh Environments

VFD AC MOTOR DRIVE INVERTER

Ideal for clean operating environments where the elimination of acoustic noise & dust disturbance is critical

» Sealed Enclosure, IP66/67/68 Ingress Protection Grade » 800 W peak power (up to 10 s) » Single 24, 48 VDC voltages

WIND TURBINE INVERTER

» Form Factor: 4.92 x 9.86 x 2.36 in. (125.0 x 250.5 x 60.0 mm)

» Low standby power consumption (< 0.35 W) » Input inrush current limiting <30 A » -SL option offers 5 VDC standby output » Remote On / Off Signal INVERTER APPLICATION TOPOLOGIES FOR AN EV WITH A DC MOTOR, A VFD-POWERED AC MOTOR, AND FOR A WIND TURBINE.

2 • 2021

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 sales@psbel.com belfuse.com/power-solutions 41

 1.866.513.2839

POWER ELECTRONICS HANDBOOK would need a 50 µF film or a 500 µF aluminum electrolytic capacitor. The probable embodiment would be a single 50-µF 800-Vdc film vs two 1,000 µF 400-V aluminum electrolytic capacitors in series. These rules of thumb may need to be tripled or more for single-phase high-impedance inputs or for large ride-through requirements. To examine the per-unit capacitance of this arrangement we can rearrange the base equations. But we shouldn’t use the capacitor voltage and ripple current; rather, we should use the motor line voltage and full-load current. Suppose the 10-hp motor is driven with 460 V and 12.4 A. Using a three-phase base power of √3VLINEILINE = 9,880 VA results in per-unit capacitance values of Cpu=3.36 for the electrolytic and 0.336 for the film capacitor. Capacitor lifetime and failure rates are exponential functions of temperature and thus of ripple current. Consequently, the ripple current stress on the dc link capacitor is critical and must be managed carefully and conservatively. With the minimum capacitance and voltage rating chosen as discussed above, the next step is to calculate the total ripple current. A rule of thumb is to choose a capacitor whose rated ripple current at high-temperature, short-duration life-test conditions is in the ballpark of the total calculated dc link ripple current. The rated “load test” current often is accompanied by tables of so-called “ripple multipliers” that apply for higher application frequencies or lower ambient temperature and derated dc voltage. But be aware that applying these multipliers to the rated ripple current reduces the capacitor lifetime to its nominal test duration, which is typically only 5 to 10 thousand hours. So, it’s a better first approximation to start with candidates whose nominal ripple current rating is close to the actual application ripple current, at least until you can perform thermal and lifetime calculations. The thermal analysis proceeds by partitioning the ripple current into two frequency bins per equation (1). The first bin constitutes the lower frequencies at the appropriate multiple of the mains frequency (depending upon the number of phases and upon the rectification or chopping scheme). The higher frequency bin is at the inverter switching frequency per equation (3) if a balanced three-phase PWM inverter scheme is applicable. Otherwise, the inverter input current and dc link current must be calculated or modeled.

SINUSOIDAL PWM, M=1

SINUSOIDAL PWM, M>1

SINUSOIDAL PWM, M<1

CURRENT FLOW

CURRENT FLOW DIAGRAM OF A THREE-PHASE VOLTAGE SOURCE INVERTER AT THE DC LINK CAPACITOR NODE. I SOURCE IS CURRENT FROM THE SOURCE ENERGY SUCH AS A BATTERY OR— IN THIS CASE— RECTIFIED MAINS, WHILE I INVERTER IS THE PULSED DC CURRENT INTO THE INVERTER. I CAP IS THE CAPACITOR AC RIPPLE CURRENT.

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EXAMPLES OF CARRIER-BASED SINUSOIDAL PWM. THE SINUSOIDAL WAVEFORM IS A REFERENCE SIGNAL FOR THE DESIRED OUTPUT CURRENT WHILE THE TRIANGLE WAVE IS THE CARRIER WITH A REPETITION FREQUENCY EQUAL TO THE INVERTER SWITCHING FREQUENCY. THE BLUE PULSED WAVEFORM IS THE PWM OUTPUT SIGNAL. FROM TOP TO BOTTOM, MODULATION INDEX M=1, OVERMODULATION M > 1, AND UNDER MODULATION M<1.

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LINK CAPACITORS

VOLTAGE AND CURRENT IN THE TIME DOMAIN

TIME-DOMAIN WAVEFORMS FOR THE THREE-PHASE INVERTER. (TO EASE VIEWING, WE USE A FAIRLY LOW 1- K H Z CARRIER FREQUENCY AND DRIVE AN INDUCTOR REPRESENTING A SINGLE-PHASE MOTOR LOAD AT 40 H Z .) THE CAPACITOR MAY NEED TO BE BEEFED UP AS THE CURRENT PULSES DRAWN FROM THE ENERGY SOURCE HAVE A HIGH AMPLITUDE. TOTAL TIME SCALE IS 50 MSEC, TWICE THE 40 H Z MOTOR DRIVE PERIOD AND 2.5× THE 50 H Z MAINS PERIOD. BELOW, THE CURRENTS OF THE SOURCE, CAPACITOR, AND INVERTER.

This method of ripple current analysis should be inherently somewhat conservative for two reasons. First, there can be some cancellation between ISOURCE, AC, RMS, and IINVERTER, AC, Consequently, ICAP, RMS may not be fully equal to the RSS RMS. (root sum of squares) value of these two ripple components. The second reason for conservatism is that the capacitor ESR generally drops with rising frequency because its dielectric loss component is proportional to the capacitive reactance. This analysis method proposes that the fundamental components of these two contributions be used in the thermal analysis, so the ESR estimate should be slightly higher than in actual practice.

PWM INVERTER PER-UNIT DC LINK CAPACITOR RIPPLE CURRENT

REFERENCES APPLICATION GUIDES FOR ALUMINUM ELECTROLYTIC AND POWER HTTPS://WWW.CDE.COM/TECH-CENTER/ FILM CAPACITORS APPLICATION-GUIDES TECHNICAL PAPERS HTTPS://WWW.CDE.COM/TECH-CENTER/ ENGINEERING-TECHNICAL-PAPERS

THREE-PHASE INVERTER CAPACITOR RIPPLE CURRENT AS A FUNCTION OF M AND φ.

CORE-TEMPERATURE, LIFETIME CALCULATORS, SPICE MODEL CODE HTTPS://WWW.CDE.COM/TECH-CENTER/LIFEGENERATORS TEMPERATURE-CALCULATORS eeworldonline.com | designworldonline.com

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POWER ELECTRONICS HANDBOOK

The advantages of Generation-Four SiC FETs NEW SILICON-CARBIDE FETS PERFORM BETTER THAN PREVIOUS VERSIONS AND CAN REPLACE SILICON MOSFETS WITH RELATIVE EASE. ANUP BHALLA , VP ENGINEERING UNITEDS i C

HOW GEN 3 AND GEN 4 STACK UP

Wide-bandgap

(WBG) semiconductors are now accepted as the future of high-efficiency power conversion. The reason: they provide lower conduction and switching losses than otherwise-comparable silicon-based IGBTs or MOSFETs. Since the introduction of WBG silicon carbide (SiC) JFETs in 2008 and SiC MOSFETs in 2011, yield, performance and costs have all improved such that it is reasonable to say overall system cost with SiC is lower than that of silicon, if benefits are fully exploited. Some inconveniences have remained though. SiC JFETs are normally-on devices, and SiC MOSFETs require particular gate drive conditions for best performance. The situation is different, however, for SiC FETS. These devices are formed from a cascode arrangement of a SiC JFET and low-voltage silicon MOSFET. They address the inconveniences of conventional SiC devices and exhibit lower losses. Generation-three SiC FETS excel in relatively high-power applications. Lowloss 650-V, 1,200-V and 1,700-V parts are becoming key enablers for high-efficiency power conversion in EVs, chargers, alternative energy, circuit protection and IT infrastructure. Gen-3 SiC FETs can displace IGBTs and the best of the available silicon ‘Superjunction’ MOSFETs at these voltage levels. Even higher voltage Technology Generation ratings can be realized with stacked arrangements, and higher Vdsmax currents are possible when devices are paralleled. Gen-3 SiC FETs now provide lowest-in-class RDS(ON) figures for 650-V and Typical Ron at RT 1,200-V devices, at less than 7 mΩ and 9 mΩ, respectively. Ron (175°C)/Ron (25°C) Development of SiC FET technology has continued beyond Qrr at 400V, 1400A/us Gen 3, driven by the need for higher efficiency and power density, along with better thermal and electrical design margins. Etot at 400V, 55A, RT, HB These goals can be realized by further reducing conduction and Passed short-circuit time at RT

A COMPARISON OF SIC FET GEN 3/GEN 4: EXAMPLE PERFORMANCE DATA FOR A SIC FET DEVICE THAT MIGHT BE USED ON A 500-V BUS.

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Passed 10us surge current of body diode at RT RthJC (°C/W) typ/max Relative die size

6mohm, 750V

7mohm, 650V

Gen 4

Gen 3

750V

650V

6mΩ

6.7mΩ

2.08

1.6

462nC

840nC

560μJ

840μJ

8μs

3μs

1570A

773A

0.21/0.27

0.15/0.19

0.65

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NEXT-GENERATION SiC COMPARING FIGURES OF MERIT FOR GEN 4 AND 5

COMPARISON OF R DS(ON) XE OSS FOM BETWEEN A SIC FET AND SOME COMPARABLE SIC MOSFETS: THE SIC FET FOM AT 25°C AND 125°C CLEARLY INDICATES THE ADVANTAGE THE SI FETS PROVIDE.

THE SIC FET GEN 4 Q RR IS LOWER THAN THAT FOR GEN 3 AND VARIES LITTLE WITH TEMPERATURE.

switching losses, but designers also want to see lower costs without a compromise in quality. Responding to such demands, latest-generation SiC FETs have better specifications. These Generation 4 SiC FETs are rated at 750 V, a significant operating voltage margin over 650-V SiC MOSFETs. Power conversion efficiency improvements arise from advanced wafer-thinning and cell density maximization processes that improve the Figure Of Merit (FoM) of onresistance/unit die area (RDS.A). This FoM combines a measure of static and dynamic losses stemming from the die size. It includes associated device capacitances and effects of reverse recovery charge Qrr. A low RDS.A value implies a high yield from wafers and potentially lower costs. In practice, the low absolute value of on-resistance doesn’t compromise current ratings. The high thermal conductivity SiC substrate and an advanced silversintered die-attach technique combine to provide good heat transfer. These benefits come without losing the easy 0-12-V gate drive characteristic of SiC FETs.

EFFECTS OF C OSS(TR)

LOW C OSS(TR) ENABLES A HIGHER USABLE SWITCHING FREQUENCY IN SIC FETS.

GEN 3/GEN 4 COMPARISON Parameters that directly affect efficiency for a SiC FET device, such as RDS(ON), Qrr and EOSS, are all significantly better with the RDS.A FoM of Gen 4. Gen 4 devices show a higher rate of increase of on-resistance with temperature, but this effect is swamped out by overall efficiency improvements compared with alternative technologies such as SiC MOSFETs. Moreover, Gen 4 devices exhibit more than double the short-circuit withstand time from eeworldonline.com | designworldonline.com

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POWER ELECTRONICS HANDBOOK THE FOM RDS(ON)XCOSS(TR) OF A SIC FET AND AVAILABLE SIC MOSFETS IN THE SAME CLASS.

AN F O M FOR GEN 3 AND GEN 4

3 µsec to 8 µsec. Body diode surge current withstand is also about double at 1,570 A for the devices considered. Another FoM, RDS(ON)xEOSS, is also a useful measure in hard-switching applications such as a Totem Pole PFC stage or a standard twolevel inverter. In these circuits, device output capacitance COSS discharges rapidly from a high voltage, producing potentially high transient power dissipation. Gen 4 devices can be designed to keep COSS and resultant stored energy EOSS low, but typically at the expense of die on-resistance and consequent conduction loss. So the FoM RDS(ON)xEOSS captures the compromise.

In hard-switching applications, it is important to keep reverse-recovery energy in any body diode effect low to maintain high efficiency. This energy varies at a low rate with temperature in Gen 4 devices. When devices are in reverse or “third quadrant” conduction, voltage drop is important as well. In SiC FETs the value is lower than with SiC MOSFETs. In a SiC FET, the drop is the sum of the JFET channel voltage, conducting in reverse at around 1.3 V, and the “knee” voltage of the body diode of the co-packaged Si MOSFET. Because the Si MOSFET is a low-voltage type, knee voltage is around 0.7 V, making the total 2 V. The comparable figure in a SiC MOSFET is around 3 to 5 V, so the MOSFET would dissipate proportionally more energy. Soft-switching applications such as LLC and PSFB converters also benefit from use of SiC FETs. Peak currents can be high in these circuits, and the low RDSON value keeps conduction losses low. Additionally, the low output capacitance COSS(tr) value of SiC FETs enables the use of higher switching frequencies because of the shorter switch turn-off delay.

HOW GEN 3 AND GEN 4 COMPARE Lowest RDS(ON) per unit area across useful temperature

Superior for high-frequency soft-switching

Low dead-time losses with superior integral diode Excellent hard-switching efficiency

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It is important in many applications to maximize switching frequency to get the ancillary benefits of smaller passive components, particularly magnetics. A useful FoM for soft-switching performance is RDS(ON)xCOSS(tr). A typical SiC FET compares well with otherwise-comparable SiC MOSFETs in this metric. Gen 4 SiC FETs clearly have an efficiency edge over comparable SiC MOSFETs, but not at the expense of ease-of-use. The latest generation parts can still be driven effectively with a 0-12-V gate signal at a maximum of ±20 V, with ESD clamp diodes incorporated. The gate threshold is around 5 V and is nominally temperature independent, unlike in SiC MOSFETs. The level of SiC FET gate drives is compatible with that for traditional Si MOSFETs or IGBTs, simplifying the task of retrofitting SiC FETs into older designs for a boost in performance. If designers optimize gate resistors and reduce snubbers in these legacy designs, swapping in SiC FETs can yield even higher efficiencies and lower costs. Even the losses in the SiC FET gate drive circuit are substantially lower, dropping from what can be watts in an IGBT circuit to near zero. Unlike in the older switch technologies, SiC FETs effectively have no Miller effect, which avoids problems of phantom turn-on as drain voltage rises. Similarly, SiC FETs are available with a Kelvin source connection which prevents source package connection inductance from interacting with the gate-drive loop, producing a similar unwanted spurious turn-on effect. With the prospect of better efficiency, higher power density and lower system costs, 750-V Gen 4 SiC FETs are a compelling choice for applications with 400 or 500-V bus voltages. They provide an enhanced margin over commonly used 650-V-rated devices in other technologies. Standardized packaging with Kelvin connection options and an advanced thermal design make the parts easy to implement into new and legacy power conversion products in traditional and emerging applications.

REFERENCES UNITED SILICON CARBIDE INC. UNITEDSIC.COM/

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CALL FOR NOMINATIONS:

THE 2021 R&D 100 AWARDS What is the R&D 100 awards program? Established in 1962, the R&D 100 Awards is the only S&T (science and technology) awards competition that recognizes new commercial products, technologies and materials for their technological significance that are available for sale or license. There are six categories in the R&D 100, listed below. There are also five special recognition categories, which follow. A given innovation can be entered in both a regular category and any of the special recognition categories — but please note that a separate entry fee is required for each nomination. Special recognition categories are awarded separately from the 100 winners that comprise the R&D 100. In addition, the judging panel will award finalist designations to selected top nominations. This announcement of finalists is made first, followed by the actual R&D 100 winners several weeks later. This allows all finalists and winners plenty of time to make arrangements to attend the awards banquet and/or conference.

SUBMIT YOUR ENTRY Starting on February 1, 2021 Deadline for submissions May 7, 2021 Late deadline for submissions May 31, 2021 To be eligible for R&D 100 Awards consideration, your product or service must have been made available to the marketplace between January 1, 2020 and March 31, 2021.

NE2W 02 1!

FOR

NEW Special Recognition Category: Battling COVID-19 This award is designed to highlight any innovation that was employed to battle the worldwide COVID-19 pandemic.

Categories include: Analytical/Test

• IT/Electrical • Mechanical/Materials • Process/Prototyping • Software/Services • Other

Special Recognitions: Corporate Social Responsibility

• Green Tech • Market Disruptor – Products • Market Disruptor – Service • Battling COVID-19

FOR MORE INFORMATION OR TO SUBMIT YOUR ENTRY, GO TO:

WWW.RD100CONFERENCE.COM/SUBMISSIONS-FOR-2021

POWER ELECTRONICS HANDBOOK

Fundamentals of fast chargers for phones and notebook PCs OVERLAPPING PROTOCOLS AND COMPETING STANDARDS HAVE CREATED A CONFUSING LANDSCAPE FOR USB CHARGERS. HERE ARE THE IMPORTANT DIFFERENCES AND SIMILARITIES OF MOST INTEREST TO DESIGNERS. MASASHI NOGAWA STAFF SYSTEMS ENGINEER QORVO

The

ﬁrst standard for charging via USB connections allowed power delivery at a 7.5-W maximum. Recently the USB standard jumped up to support 100 W. The fast charging this change makes possible is attractive but can be confusing for designers to navigate. Several charging protocols have been developed to handle the new capabilities, and programmable power supplies (PPS) now help mitigate the heating challenges.

THE CC-CV CHARGING CYCLE

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To understand the advantages of this generation’s fast charging technologies, it is helpful to review the battery charger fundamentals of the lithium-ion packs powering our mobile devices. On a charger, greater currents and higher voltages charge batteries faster, but too much power leads to overheating, and we all know what happens when a phone battery gets too hot. To prevent overheating, the charger circuit uses a charge controller IC to regulate the ﬂow of battery current using a constant-current to constant-voltage (or CCCV) charting algorithm. This technique is the key to battery charging technologies. CC-CV charging happens in two phases. In the constant-current phase battery voltage is much lower than its target. The fast charger pushes high currents into the battery, speeding the charging process. The constant-voltage phase begins once the battery has received most of its charge. At that point, the IC will reduce the current to prevent overcharging. Heat is among the most signiﬁcant challenges designers face when developing battery chargers. An entire industry has emerged to help designers walk the narrow boundary between faster charging time and safe temperatures. Most battery chargers now monitor multiple points of temperature for safety. If they detect a high temperature, a controller reacts immediately to reduce the

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FAST CHARGING

FAST CHARGER WITH PPS

A TRADITIONAL BATTERY CHARGING SYSTEM USING THE STANDARD TYPE-A CABLE (TOP) AND A NEW USB-C BASE PD3.0 PPS SYSTEM (BOTTOM). THE TRADITIONAL TYPE-A CHARGER HAS TWO HEAT SOURCES THE BATTERY PACK AND THE CHARGER FET FROM DC-TO-DC CONVERSION INSIDE THE PHONE/TABLET. THE PPS CHARGER SYSTEM CONVERTS DC-TO-DC AT THE CHARGER SIDE OF THE C CABLE TO ELIMINATE ON HEAT SOURCE.

charging current or voltage. There have also been attempts to establish industry standards like those of JEITA (Japan Electronics and Information Technology Industries Association) to promote safe best practices. The major sources of overheating are the lithium-ion battery packs and the transistors of CC-CV charger ICs. Not much can be done about the battery pack heat-after all, the battery must charge. Instead, designers focus on dissipating the heat from the charger ICs. Consider a traditional battery charging system using the standard Type-A USB cable. This setup must mitigate heat from both the charger and the battery. The USB-A delivers 5V (and 9 or 12V in some propriety schemes) to the device using the VBUS wire, but a lithium-ion battery can take a maximum 4.2 V. Thus the CC-CV controller in the Charger IC block must reduce the voltage input to 4.2 V for charging. A power loss always accompanies the conversion of one dc voltage to another, generating unwanted heat. Thanks to technological innovations, we can remove the CC-CV controller from mobile devices. Thus the battery pack is now the principal heat source. The basic structure of a Fast Charger that uses the latest USB-C cable and PPS protocol uses a power delivery controller, or PD-

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Controller. The PD-controller coordinates with the PPS charger to pass voltage from the USB-C (VBUS) directly to the battery without dc-to-dc conversion. This approach dramatically reduces the power losses that cause heat. The PD-Controller + PPS Charger block also monitors battery charging voltage, current, and temperature to optimize CC-CV charging. It determines target values for voltage and current, then requests the right levels from the charger at the other end of the USB-C cable. This approach gives designers precise control over output current and voltage. For example, Qorvo PPS power ICs allow for voltages from 3.3 to 21 V in 20-mV steps and currents from 1 A to a charger’s maximum in 50-mA steps. In another example of a Fast Charger implementation with PPS, VBUS of the USB-C cable directly goes to the battery pack(s). Higher charging current flows over the entire charging path, causing higher conduction power loss (= Icharge2× Rpath). With recent dc-to-dc voltage conversion technologies, high-efficiency switching takes place at a specific input and output voltage ratio. For example, a ratio of Voutput = Vinput × ½ can be efficient. When VBUS voltage doubles, both the charging current and conduction losses are halved.

2 • 2021

THE USB IMPLEMENTERS FORUM FAST CHARGER AND CHARGER LOGOS. AN EXAMPLE IS QUALCOMM’S QC4+/QC4 CHARGERS BASED ON THE USB PD 3.0 PPS SPEC.

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POWER ELECTRONICS HANDBOOK FAST CHARGING PROTOCOLS OEMs like Apple, Samsung and Xiaomi have been increasingly adopting fast charging features, while new technologies like USB Power Delivery and Qualcomm Quick Charge are competing for market share. And power engineers expect many USB chargers will support the Power Delivery 3.0 Programmable Power Supply (USB PD 3.0 PPS) standard. It is worth taking a closer look at recent fast charging device applications and the competing standards. There is a special USB Implementers Forum (USB-IF) Fast Charger logo to indicate a charger is certiﬁed for 80 W USB 3.0 PPS. There is also a “non-fast” charger logo. The only difference is the lower red line of the logos read “FAST CHARGER” or just “CHARGER” respectively. Several phones employ this USB Fast Charger technology. For example, Samsung calls it Super Fast Charging or Super Fast Charging 2.0 on its USB PD 3.0 PPS-supported Note 10+ and Galaxy S20 5G devices. Xiaomi phones that include Mi 8 and Mi 9 support QC4+. Now consider how actual charging voltage and charging current change over time on a modern smartphone. An Xiaomi Mi 9 phone is charged by a QC4+ charger, ACT4751M_101_REF05. Throughout the entire charging sequence, this Xiaomi phone maintains its maximum charging current exactly at 2.9 A though the charger supports 4 A and the USB-C cable can handle 5 A. A PD controller clips charging current peaks at 2.9 A in this Xiaomi phone. At the beginning of this charge sequence, the phone demands around 9.6 V of VBUS voltage and, eventually, the phone demands 7.0 V by reducing VBUS voltage stepby-step over about 2.5 minutes. Once VBUS reaches 7.0 V, a charger IC inside the phone maintains

operation at this voltage. It appears this 7.0-V supply point is an optimal operating point for the Mi 9 charger IC. Thus the Xiaomi Mi 9 uses the USB PD 3.0 PPS function and/or QC4+ function to program VBUS at 7.0 V, different than the standard 5/9/15/20-V options.

TYPE-A CABLE During the era of Type-A port charging, many vendors adopted features (some incompatible with today’s standards) to support higher power and faster charging. These Type-A port technologies remain in USB-C protocols today for the sake of backward compatibility. The original Type-A port specification supported only 2.5 W (5 V/0.5 A). These days, it is difficult to find a charger supplying so little power. The original 2.5-W USB charger spec was upgraded to 7.5 W by the specification of Battery Charge rev. 1 (BC1.2). This remains the highest official power level USB-IF recognizes for Type A ports. All other higher power Type-A port protocols are not compatible with USB-IF standards. There are many privately defined charging methods available that are outside of official USB specifications. Examples include Qualcomm’s Quick Charge 2.0 and Quick Charge 3.0, along with protocols for phones from major OEMs like Apple, Samsung, and others. At a Type-A port, there are only four pins available--one pair of VBUS and GND (power rails) and D+/D- (data lines). To charge a device through a Type-A port, we need VBUS and GND constantly, so all Type-A port base charging methods use D+/D- lines for logic handshaking to establish charging control. Protocols including BC1.2, Qualcomm QC2 and QC3, and those used by Apple and Samsung all employ the D+/D- lines in different ways for handshaking. The details of these protocols are beyond the scope of this article, but it is important to note we use D+/D-

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2 • 2021

AN EXAMPLE OF A FAST CHARGE SEQUENCE FROM A MODERN PHONE. FOR THIS DEMO, AN XIAOMI MI 9 PHONE CHARGED FOR 50 MINUTES FROM A VOLTAGE REGULATOR DESIGNED FOR USB PD CONTROLLERS (QORVO ACT4751M_101_REF05). THE PHONE CONNECTS TO THE REGULATOR THROUGH A USB-C CABLE WHILE AN ADAPTER FOR RECORDING THE POWER DELIVERY PROTOCOL TRAFFIC (FROM TOTAL PHASE) READS OUT THE CHARGING PROFILE, STARTING WITH THE BATTERY 40% CHARGED. BELOW, AN ENLARGED VIEW OF THE FIRST FOUR MINUTES OF THE CHARGING SEQUENCE.

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FAST CHARGING

USB PROTOCOL TESTERS CAN TELL WHETHER OR NOT A USB-C PORT SUPPORTS PPS. HERE, A VOLTAGE REGULATOR (ACT4751M_101_ REF05) HAS BEEN CONFIGURED TO PROVIDE POWER WITH AND WITHOUT PPS FOR USB CHARGING. A USB POWER METER LOAD TESTER (AVHZY CT-2) MONITORS THE CHARGING BEHAVIOR. THE TESTER READOUT (TOP) SHOWS THAT BOTH PORTS SUPPORT FOUR FIXED VOLTAGE OPTIONS (IN THE PD 3.0 DOCUMENT, CALLED PDOS) OF 5 V/3 A, 9 V/3 A, 15 V/3 A AND 20 V/4 A. THE PORT ON THE RIGHT HAS THE ADDITIONAL OPTION OF PROGRAMMABLE POWER 3.3–21 V/4 A.

lines for charging controls because USB-C ports include D+/D- lines for backward compatibility. It is worth visiting Quick Charge 3.0. Though QC3 is not a USB PD 3.0 base, it achieves the same goal of PD 3.0 PPS or QC4+/QC4 through Type-A ports. QC3 supports programmable voltages in 200 mV steps. The intention of this ﬁne-step voltage control is the same as for PPS: to reduce power loss and heat from a battery charger IC inside a phone or tablet.

Unlike the Type-A port specification, the power delivery 3.0 (PD 3.0) specification of USB-C supports up to 100 W of power delivery from the onset, so almost all USB-C chargers follow this spec. One source of confusion is that the physical specifications for the USB-C port differ from PD 3.0 specifications. This has important implications: Not all USB-C ports support PD 3.0. Some uses of physical USB-C ports replace old Type-A and Type-B connections where only D+/D- pins are used. Thus a physical USB-C port does not guarantee the existence of PD 3.0 fast charging capabilities. Many charger products with USB-C ports are normal PD chargers, not USB Fast Chargers. Based on the PD 3.0 specifications, standard VBUS voltage options are 5, 9, 15 and 20 V. Because these normal PD chargers produce a fixed (non-programmable) VBUS voltage, they are the same as traditional Type-A port chargers in terms of battery charging efficiency. One easy way to identify USB Fast Chargers with PD 3.0 PPS capabilities is the USB logo: The red bottom line will say “FAST CHARGER.” To distinguish their products from normal non-PPS PD

USB TYPE-C TO USB 2.0 MICRO-B CABLE ASSEMBLY

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A USB-C TO USB 2.0 MICRO-B CONVERTER CABLE LETS PHONES WITHOUT A USB-C PORT TO STILL QUICK-CHARGE FROM A QC4+ CAPABLE USB-C CHARGING PORT.

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chargers, most of vendors get their faster chargers certified by the USB-IF. A USB protocol tester can easily discern whether or not a USB-C port supports PPS. In addition to the basic fixed 5, 9, 15, and 20-V options, a USB Fast Charger offers a programmable output option. A charger with Quick Charge 4 can be compatible with USB Fast Charger. It is also possible to get one product certified for both USB Fast Charger and Quick Charge 4. A certification test of QC4 chargers requires tighter regulation of output voltage and current compared with that for USB Fast Chargers. Although it is difficult to tell the difference between a USB Fast Charger and QC4 charger without logos or product labels, there are some key differences. For example, QC4 chargers can report the charger’s internal temperature by its specification. This feature can potentially let a phone supporting QC4 charge its battery faster than USB Fast Chargers. The difference between QC4 and QC4+ is the Quick Charge 2.0/Quick Charge 3.0 backward compatibility. QC4+ supports QC2 and QC3 through D+/D- lines. Through use of a USB-C-toUSB-2.0 Micro-B cable, phones without a USB-C port can still get a quick charge from a QC4+ capable USB-C charging port. Because QC2 and QC3 are incompatible, a QC4+ capable charger

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cannot get certified as a USB Fast Charger. To make matters more complicated, there are some Asian phones whose connectors have the physical shape of USB-C, but their communication/handshaking protocols do not follow the USB PD 3.0 specification. Thus it is easy to see why power electronics engineers have a difficult time explaining new fast charging technologies consumers. The indistinct marketing and branding efforts for these new technologies has only added complexity.

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TOP, CONNECTOR HARDWARE USED FOR THIS TEST. (BELOW) USB-C CHARGING PORTS FROM A CHARGER (ACT4751M_101_REF05) CONNECT TO TYPE-A PORTS ON A USB POWER METER LOAD TESTER (AVHZY CT-2). ONLY THE TRADITIONAL FOUR WIRES (VBUS, GND, D+ AND D-) ARE USED. AS IS EVIDENT ON THE USB POWER METER LOAD TESTER DISPLAY, THE USB-C PORT ON THE LEFT SUPPORTS QC2 AND QC3 WHILE THE ONE AT RIGHT PROVIDES NO QC2 AND QC3 SUPPORT, ONLY BC1.2 SUPPORT.

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Bel Power Solutions ................................................ 41

Newark, An Avnet Company .................................. 22

Coilcraft ..................................................................BC

Pico Electronics ...................................................... 19

Digi-Key ......................................................Cover, IFC

RECOM Power GmbH .............................................. 9

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Ironwood Electronics .............................................. 18

Sorbothane ............................................................. 10

Keystone Electronics Corp ................................1, IBC

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