Test & Measurement Handbook 2022

Scope-based diagnosis of three-phase motor drives Page 6

JUNE 2022

5G-test equipment goes mainstream Page 26

EMI testing for IoT transceivers Page 43

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TE ST & M E AS U R E M E N T HA N DB OOK

A frequency you can count on There

are few constants in life, but what few there are might include death, taxes, and a U.S. grid frequency that

doesn’t vary by more than ±0.5 Hz. However, the certainty of the grid frequency is coming into question, thanks to the rising percentage of renewable energy sources that supply the grid with power.

The problem is that renewable sources such as solar panels and wind turbines have little in the way of inertia. Inertia in power systems refers to the kinetic energy stored mainly in large rotating generators. Inertia lets generators remain rotating despite temporary interruptions such as a glitch in the fuel supply. Energy arising from inertia can be particularly valuable as it can temporarily make up for power lost from a generator that fails. Inertia energy—typically available for a few seconds—gives power plant controls time to detect and respond to the failure.

But inertia doesn’t just keep power plants online. Inertia resists a drop in grid frequency, which is a measure of the balance of the supply of electricity and demand. Inertia basically gives the grid time to rebalance supply and demand and keep the grid frequency stable.

Inertia tends to be more important in smaller grids. Consider the grid in the United Kingdom with a peak demand of around 30 GW, a little less than half that of ERCOT (Electric Reliability Council of Texas) which is the smallest grid in the U.S. To boost its inertia, the U.K. recently installed a pair of roughly 200-ton flywheels spinning at 500 rpm. Smaller flywheels with speeds of 1,500 rpm will come on-line there later this year, and more are planned. These measures are necessary because the U.K. grid currently gets about 30% of its power from inertialess wind and sunlight.

In the U.S., the ERCOT grid has the most worries about inertia partly because of its size but also because it contains a large amount of wind energy, accounting for 23% of the state’s power. So far, ERCOT has compensated for declining inertia by adopting several inexpensive methods such as shedding fast-responding noncritical loads in response to changes in frequency.

Eventually, even managers of large grids will need to worry about inertia with the growth of renewable energy sources. A few renewable energy schemes do,in fact, have some built-in inertia. They include biomass-powered generators, concentrating solar power plants, and of course hydropower.

You might think that wind turbines would have a lot of kinetic energy in the rotating mass of their blades, shafts, and generators that could augment grid inertia. But most modern wind turbines don’t carry synchronous generators, those that inherently resist changes in frequency. Wind turbine kinetic energy may help in the future,

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though, by giving turbines a way to actively sense grid frequency so their generators can be programmed to temporarily boost output beyond that supported by steady state wind speeds.

Solar panels can also provide inertia-like properties—sort of. The idea is to operate the panel such that its output is below that available based on instantaneous weather conditions, basically so it has power in reserve. The panel inverter can then rapidly increase output to provide frequency support.

Problem is, the panel must have some reserve capacity to provide this kind of support, so there’s an economic tradeoff between providing energy and being able to respond to imbalances. On the bright side, generators that incorporate inverters can respond more quickly to frequency dips than conventional rotating generators.

The final word on inertia isn’t in. But the day may come when lab tests will have to add ac line frequency to the list of items that must be verified rather than assumed.

Leland Teschler • Executive Editor

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CONT E NT S T E S T

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M E A S U R E M E N T

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A FREQUENCY YOU CAN COUNT ON

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SCOPE-BASED DIAGNOSIS OF THREE-PHASE MOTOR DRIVES ORDINARY OSCILLOSCOPES CAN VIEW BASIC PARAMETERS IN MODERN MOTOR DRIVES, BUT OPERATIONAL INSIGHTS GENERALLY REQUIRE SPECIALIZED POWER ANALYZERS.

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MEASURING ESR AND ESL OF DC-LINK CAPACITORS IMPEDANCE ANALYZERS WORK BETTER THAN LCR METERS WHEN IT COMES TO SIZING UP THE SUPER-LOW IMPEDANCE AND INDUCTANCE OF MODERN POWER CONVERTER CAPACITORS.

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AUTOMOTIVE CABLE AND HARNESS TESTING MADE EASY

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DEBUGGING DIGITAL AND ANALOG SIGNALS WITH CROSS-DOMAIN ANALYSIS MULTI-DOMAIN SIGNAL ANALYSIS IS BLURRING THE LINE BETWEEN REAL-TIME OSCILLOSCOPES AND REAL-TIME SPECTRUM AND SIGNAL ANALYZERS.

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5G TEST EQUIPMENT GOES MAINSTREAM

AUDIO MEASUREMENTS FOR PRODUCT DEVELOPMENT

EXCEEDING 100 GBPS WITH A SUB-TERAHERTZ 6G TESTBED HERE’S A LOOK AT TESTING NECESSARY FOR SUPER-HIGH THROUGHPUTS INVOLVED IN THE EMERGING AREA OF 6G.

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MEASURING AND USING STATIC ELECTRICITY PUT AWAY THE MULTIMETER IF YOU WANT TO SIZE UP STATIC ELECTRICITY. HERE’S WHY.

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KEY CONSIDERATIONS FOR RF POWER MEASUREMENT EQUIPMENT THE CHOICE OF POWER SENSOR AND METER GREATLY DEPENDS ON THE QUALITIES OF THE SIGNAL BEING MEASURED.

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TEST AND MEASUREMENT IN EDUCATION LAB CLASSES BECOME MUCH MORE MEANINGFUL WHEN STUDENTS CAN OPERATE THEIR OWN TEST INSTRUMENTS RATHER THAN WATCH AN INSTRUCTOR TURN KNOBS AND PUSH BUTTONS.

MEASURING POWER CHOKE INDUCTANCES THE INDUCTANCE OF POWER CHOKES CHANGES WITH CURRENT LEVEL. CONVENTIONAL SMALLSIGNAL-MEASURING BRIDGES ARE OUT OF THEIR DEPTH HERE!

J U N E

IT PAYS TO KNOW THE FUNDAMENTAL PARAMETERS THAT QUANTIFY THE PERFORMANCE OF SOUND EQUIPMENT.

A HIPOT TESTER, SWITCH, AND AUTOMATED SOFTWARE CAN ASSURE VEHICLE CONNECTORS ARE SAFE AND RELIABLE.

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PERFORMANCE QUALITIES ONCE ONLY AVAILABLE IN EXOTIC INSTRUMENTS CAN NOW BE FOUND IN GEAR DESIGNED FOR PRODUCTION TESTING.

THE MANY MEASUREMENTS OF POWER QUALITY THERE’S A CONFUSING VARIETY OF WAYS TO QUANTIFY POWER CONSUMPTION. MAKE SURE YOU UNDERSTAND WHAT YOUR FAVORITE POWER METRIC MEANS.

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APPLYING PC-BASED TEST & MEASUREMENT INSTRUMENTS FOR SOME FEATURES, PC-BASED SCOPES SPORT HIGHER PERFORMANCE THAN SIMILAR BENCHTOP INSTRUMENTS.

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EMI TESTING FOR IOT TRANSCEIVERS A FEW OSCILLOSCOPE MEASUREMENTS CAN GIVE INSIGHT INTO SOURCES OF ELECTROMAGNETIC INTERFERENCE THAT GARBLE THE OPERATION OF WIRELESS IOT DEVICES.

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WHAT YOU NEED TO KNOW ABOUT ERROR ANALYSIS IN PCIE 6.0 DESIGNS THE MUCH-AWAITED NEXT GENERATION OF PCIE MAKES IT IMPORTANT TO UNDERSTAND BIT ERROR-RATE MEASUREMENTS.

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Scope-based diagnosis of three-phase motor drives Leland Teschler, Executive Editor

Variable

Ordinary oscilloscopes can view basic parameters in modern motor drives, but operational insights generally require specialized power analyzers.

PWM to vary the frequency and voltage delivered to the motor. drives The modulation on these PWM (VFD) generate carefully controlled waveforms is complex, making pulse-width modulated (PWM) it challenging to get stable oscilloscope measurements to waveforms to modulate the speed diagnose problems. of industrial ac motors. Often, the Use of PWM motor drives has motors involved are three phase. been accelerated by the perfection of vector control methods which Three-phase ac induction motors allow dc motors to be controlled (ACIMs) are widely used though with the precision and reliability they are less efficient than of ac motors. These advances let brushless dc motors (BLDC) and BLDCs and PMSMs replace brushed permanent magnet synchronous dc motors and induction motors not The typical functional blocks of a three-phase motor drive motors (PMSM). Each of these only in industrial applications but control, also called field-oriented PWM drives can be powered motor systems requires its own also in power tools, appliances, and control (FOC), where the stator by dc, single-phase ac, or threespecial drive, but all of them use electric vehicles. currents of a three-phase ac or phase ac. But three-phase Typical trapezoidal drive signals BLDC motor are identified as supplies are common in industrial two orthogonal components equipment. The three-phase visualized with a vector. supply is rectified and filtered to In contrast, scalar control produce a dc bus which powers simply changes the fundamental an inverter section inside the frequency of the PWM waveform drive. The inverter consists of driving the motor. To maintain full three pairs of semiconductor torque, the control system in the switches (MOSFETs, GTOs, drive maintains the ratio between IGBTs, and increasingly SiC the output voltage and PWM power transistors) with associated fundamental frequency. diodes. Each pair of switches Control electronics generate powers one phase of the motor. three low-frequency sinewaves This is the basic architecture, but 120° apart which modulate the the control electronics varies in terms of feedback and complexity. width of the pulses for each pair of switches. The average voltage Nevertheless, there are a few general categories of PWM drives. presented to the motor winding is basically sinusoidal. The other two The six-step/trapezoidal phases of the motor winding have drive generally powers BLDC similar average voltages spaced motors. In contrast, scalar 120° apart. drives typically handle induction In scalar control, the motor Hall sensors typically provide feedback in a simple six-step trapezoidal motors. The term scalar is used to appears as an inductor to the contrast this method with vector drive. The drive outputs U, V, and W are applied to the motor stator.

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VFD ME ASUREMENTS Typical pulse-width-modulated motor drive signal

Scope connections

Typical scalar drive signal. The average phase-to-phase voltage of the PWM waveform between two phases is sinusoidal. output voltages of the inverter. Because an inductor has a higher impedance to higher frequencies, most of the current drawn by the motor is from the lower frequency components in the PWM waveform. This results in the current drawn by the motor being approximately sinusoidal in shape. By controlling the amplitude and frequency of the modulating waveforms, and controlling the V/Hz ratio, PWM drives supply three-phase power to drive the motor at a required speed.

Advanced drives for induction motors and synchronous motors employ vector drive techniques. These drives are more flexible and efficient than scalar drives, but also more complex. Like scalar drives, vector drives power the motor with sinusoidal current. The vectors, D and Q, are orthogonal and have magnitudes related to the motor torque and magnetic flux. The controller uses what’s called the Clarke and Park transform to calculate the magnitudes of D and Q and then uses these values as setpoints for the control loop.

The control system measures the position

of the rotor to synchronize the system. Rotor sensing typically employs Hall sensors or a quadrature encoder interface (QEI). But there are also sensorless systems where the control system uses the back-emf of the motor to determine rotor position.

Three-phase voltage relationships The ideal three-phase voltage signals.

Typical vector control system block diagram

Clarke and Park DQZ (direct quadrature zero) transforms are commonly used in fieldoriented control of three-phase ac machines. The Clarke transform converts the time domain components of a three-phase system to two components in an orthogonal stationary frame.

Within the control system, these transformations are used to convert the three-phase voltages being applied a motor to orthogonal D and Q vectors. eeworldonline.com

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VFD measurements make use of both voltage and current probes. Motor drives involve relatively high voltages. For example, the dc bus voltage in a 480-Vac three-phase motor drive is typically around 680 Vdc. Most scope probes aren’t rated for voltages this high.

Common-mode voltages can also be relatively high. That is, measurements are often “floating” relative to ground, so groundreferenced probes may be dangerous. It is important to verify signals are not floating more than the probe’s common-mode voltage rating. Most VFD frequencies of interest are 200 MHz or less, a factor to keep in mind when considering probe bandwidth. For these reasons, scope makers generally recommend high-voltage differential probes for generalpurpose use in VFD measurements.

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A point to note is that ground-referenced passive probes should not be used to measure phase-to-neutral voltages. The neutral terminal on a power feed is probably not at ground potential. So significant currents will flow through the probe and oscilloscope earth ground. The result can be shock or damage to the DUT or scope.

Before any power measurements, current probes must be degaussed, and all probes should be de-skewed. Residual magnetization can throw off measurements. Degaussing removes any residual magnetization from the probe’s magnetic core. The degaussing procedure typically commences with just a button press.

The de-skewing process corrects for differing propagation delays arising between any two scope channels as can be introduced DESIGN WORLD — EE NETWORK

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TE ST & M E AS U R E M E N T HA N DB OOK Single-phase three-wire system

Single-phase, three-wire wiring is rare in industrial settings but common in consumer and light commercial installations. by probes and probe cabling.

De-skewing is important because phase relationships are critical for many of the measurements on VFD systems. The basic procedure is to put a synchronized signal on all channels and make adjustments to equalize the delays in individual

phase VFD connections. The most straightforward is the single-phase two-wire connection. Two scope channels, one for voltage and the other for current, are necessary.

Single-phase ac and dc systems use the same setup.

The second type of singlephase VFD connection is singlephase three-wire. It is rare in industrial uses but sometimes arises in North American residential applications where one 240-V and two 120-V supplies are available and may have different loads on each leg. VFD measurements require a four-channel scope, two voltage channels and two current channels. The total power measured is VxI (Load1 + Load2).

We now turn to measuring three-phase connections with four scope channels, two for voltage and two for current. The output of

Three-phase, three-wire, two power measurements

Here currents are measured on phases A and B, and voltages are measured on phases A and B with respect to phase C. The total true power is then the instantaneous power VAC x IA plus VBC x IB. channels. Scope makers such as Tektronix offer special de-skewing fixtures to simplify the process.

Though three-phase VFDs are perhaps the most common type, commercial, residential, or automotive drive systems may be powered by single-phase ac or dc. In addition, threephase systems can be wired and modeled in two ways: star (or wye) and delta connections. The wiring configuration determines the calculations for the power analysis. These configurations apply to both the inputs and outputs of motor drives.

There are two kinds of single-

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a three-phase VFD is typically three wires. Motor drive inputs are more likely to use a four-wire system.

When three wires connect the source to the load, test setups use one of the wires as the reference phase. Then the scope is set up to measure voltage between the reference phase and each of the other two. And a current measurement is made on each of the two non-reference connections.

The load and source can be wired in delta or star configurations but there must not be a neutral conductor between them. In this situation two current and voltage readings will account for the total

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Three-phase, measurements with six scope channels

When there is no physical neutral, it is possible to determine the instantaneous line-to-neutral voltages from the instantaneous line-to-line voltages: VAN= (VAB-VCA)/3;VBN=(VBC-VAB)/3;VCN=(VCA-VBC)3 power being delivered to the load.

Though only two power readings--one on each nonreference phase--are necessary to measure the total power in a threewire system, there are advantages to using three. The setup requires six oscilloscope channels: three for voltages and three for currents.

This configuration provides individual phase-to-neutral voltages and the power in each individual phase, something not possible when using one of the connections as a reference.

Although there is no physical neutral in this system, it is possible to determine the instantaneous line-to-neutral voltages from the instantaneous line-to-line voltages. The line-to-neutral voltage for a given line is just the difference between the voltage measured to each of the other two lines, divided by three.

The resulting point-by-point

conversion expresses all voltages relative to a single reference and corrects the phase relationships between voltage and current for each phase. This conversion also allows instantaneous power calculations--total true power supplied to the load is just the sum of the line-to-neutral voltage multiplied by the current in the line for each of the three lines.

That brings us to the case where there are three phase connections to the load along with a neutral connection. Here, measurements take place via six scope channels, three for current and three for voltage. Line voltages are all measured relative to the neutral. Phase-to-phase voltages can be calculated from the phaseto-neutral voltage amplitudes and phases using vector mathematics.

The total power is the sum of the power measured for each of the three phases.

Power measurement with six scope channels

It takes six channels to measure the total power in a system employing a neutral conductor between the line and the drive or between the drive and the motor. Voltages are all measured relative to the neutral. eeworldonline.com

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VFD ME ASUREMENTS Phasors Of course, the three sinusoidal ac voltages and currents in a three-phase motor drive all have the identical frequency. Having six or more waveforms plotted on the same time axis makes it difficult to discern magnitudes and phase angles. That’s why phasor diagrams representing the magnitude and directional relationship between two or more vectors are widely used in threephase power system analysis.

It’s complicated to set up a threephase phasor diagram on ordinary scopes, but instruments targeting power quality measurements have phasordiagram software built in.

Perhaps the most complicated part of the measurement procedure is in orienting the three or more clamp-on current probes normally used in power quality measurements—they must be connected so arrows on the inductive head point toward the load. Motor loads are inductive, so the voltage vector on the phasor diagram should lead the current vector. If the current vector leads, the current clamp is probably reversed.

In a balanced three-phase system, with identical loads on each phase, each of the three voltage vectors nominally sit 120º apart from each other. Ditto for the current vectors. In this balanced system the sum of the line currents is zero at any instant; the same for the sum of the line-to-line voltages. But real systems have differences in load impedances among the three phases that result in imbalances. The resulting phasor diagram contains vectors of different lengths and different angles between voltage and current vectors.

The phasor diagram is helpful for seeing the impact of inductive or capacitive loads. Phasors for pure resistive loads have in-phase voltage and current, meaning there is no lag between voltage and current. Motors are inductive in nature, so the current vector always lags the corresponding voltage vector.

Motor drive designers strive to minimize the phase angle between voltage and current phasors.

Clarke and Park DQZ (direct quadrature zero) transforms are commonly used in field-oriented control of three-phase ac machines. The Clarke eeworldonline.com

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transform converts the time domain components of a three-phase system to two components in an orthogonal stationary frame. Within the control system, these transformations are used to convert the three-phase voltages being applied a motor to orthogonal D and Q vectors. Basically, the DQZ transformation rotates the reference frame of a threeelement vector to simplify the analysis.

These simplified vectors can easily be scaled and integrated to maintain a desired speed. Reverse transforms may then be used to create the drive signals for PWM in the inverter.

The D and Q vectors generally can’t be directly measured because they typically reside within a digital signal processing block such as an FPGA. But power analysis instrumentation often contains software that can derive measurements of D and Q based on the three-phase output voltage or current.

This capacity enables helps gauge the effect of control system adjustments.

The output waveform of a PWM drive is a complicated mixture of highfrequency components related to the carrier and components at lower frequency related to the fundamental frequency driving the motor.

Setting up a trigger for scope measurements on PWM waveforms can be challenging because the waveform is modulated at low frequency.

Measurements such as total rms voltage, total power and so forth must take place at high frequency but over an integral number of cycles of the low frequency component in the output waveform.

Consequently, specialized power analyzers often contain facilities for simplifying these measurements. In the case of Tektronix instruments, for example, software demodulates the PWM waveform on a channel specified as an “edge qualifier” and extracts the envelope as what’s called a math channel.

A phasor diagram shows the relationship between voltage and current for each phase. Specialized power analyzers generate phasor diagrams automatically.

This example is from a Tektronix instrument.

References Tektronix, www.tek.com Keysight Technologies, www.keysight.com

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Applying PC-based test & measurement instruments For some features, PC-based scopes sport higher performance than similar benchtop instruments.

Trevor Smith, Pico Technology

The

debugging and validation of electronic systems calls for versatile

instruments with wide-ranging functions and easy programmability. An engineer working on high-speed logic signals one day might be called upon to run a long-duration soak test the following day and check for immunity to power harmonics the next.

A high-performance PC-based scope: The PicoScope 6000E series sports -3 dB bandwidths of up to 1 GHz and a capture memory (shared between active channels) of up to 2 GS. Some models also handle a maximum single capture duration of up to 1.6 sec at the max sampling rate.

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PC-based test instruments such as those from Pico Technology address such needs.

Instrument acquisition hardware sits in a compact external enclosure that connects to the host computer, typically with a USB 2.0 or 3.0 connection. The enclosure hosts connectors for the input and output signals to and from the device-under-test (DUT) and a separate power connection, where required.

This sort of economical PC-based instrumentation is possible thanks to advanced FPGA technology that controls the channel settings, signal conditioning, triggering, on-board memory management and communication with the host PC. The FPGA does the heavy-duty processing of test data acquisition and/or signal generation.

Data flowing on the USB interface is limited to commands, responses and composite information required for processing and display of graphical or numerical test data.

Separation of the acquisition hardware from the analysis and display platform allows Pico T&M products to benefit from improvements to PC and display technology as it becomes available.

Pico instrument software works with any Windows 10 or 11 machine and other industry standard platforms. Most Pico software packages can take advantage of high6 • 2022

definition and UHD displays for presentation of waveforms, measurements and analysis results.

It may be useful to review features of widely available PC-based test instruments. PicoScope oscilloscopes feature real-time bandwidths from 10 MHz to 1 GHz and a choice of two, four or eight analog channels, and up to 16 digital channels on the MSO models. There are flexible and high-resolution models too, with up to 16 bits for precision analog measurements. Most models have an integrated Function Generator/AWG. The next generation PicoScope 7 user interface, with versions for Windows, Linux & Mac operating systems, delivers six instrument functions: oscilloscope, spectrum analyzer, serial protocol analyzer (30 serial decoder/analyzers included, with more in development), logic analyzer (on MSO models), and a function/sweep generator.

Most T&M PicoScope models include a signal generator that can deliver sine, square, triangle, ramp, sin(x)/x, Gaussian, half sine, white noise and PRBS waveforms. This is complemented with a sweep function that enables device testing over a predefined frequency range. This sweeping can be helpful in testing filters, operational amplifiers, etc., over their specified operating range.

More complex devices need testing with real-world waveforms, which is where an AWG comes in handy. Many PicoScopes include an AWG, which can now be fully controlled from within the PicoScope 7 user interface.

Real-world waveforms captured with a PicoScope can be loaded into the AWG memory and replayed as needed for device testing and performance verification. For an example of where this capability can be useful, consider a staged car crash with sensors set to record the impact characteristics. It’s an expensive test, and something that you would only want to do once. But the ability to save the waveforms allows them to be subsequently replayed as often as needed with the PicoScope AWG for development of safety-related systems.

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PC -BASED S CO PES

Serial decoding is a standard feature of PicoScope instruments. The decoded data can be displayed in graph mode as depicted here, in tabular format, or in both formats simultaneously. Graph format shows decoded data in a bus format, aligned with the analog waveform, on a common time axis, with error frames marked in red. Frames can be zoomed and correlated with acquired analog channels to investigate timing errors or other signal integrity issues that are a root cause of data errors. modified with the AWG editor to stress the DUT and see how it behaves at the extremes of its specification, and with ‘known good’ and ‘known bad’ signals for go/no-go testing.

Also of note is that the decoding of serial protocols is included in PicoScope as standard. The decoded data can be displayed in graphs or tables or in both of these ways simultaneously. Serial protocols have become particularly important in vehicles because the growing number of vehicle convenience and comfort features have led to the demand for lower-cost wiring alternatives.

LIN (Local Interconnect Network) was developed to provide a common sensor/ actuator bus standard. Initially introduced in 2000 as an inexpensive serial communication bus, the latest version (2.2A) is now standardized as ISO 17987. LIN uses a simple and low-cost single wire physical layer implementation based on ISO 9141 (as used by the K-line onboard diagnostic standard).

This single-wire implementation does make LIN more susceptible to EMC than two-wire buses, which in turn limits the data rate to 20 kb/sec and the recommended number of nodes to 16.

Although serial buses like LIN offer several advantages, they also present difficulties during troubleshooting and debugging because data transmits in packets or frames that must be decoded before the designer can make sense of the information flow. Manually decoding (or “bit counting”) streams of binary data is error prone and time-consuming.

PicoScope includes decoding and analysis of widely used serial standards to help engineers see what is happening in their design to identify programming and timing errors and check for other signal integrity issues. Timing analysis tools help to show performance of each design element, enabling the engineer to identify those parts of the design that need to be improved to optimize overall system performance. Frames can be zoomed and correlated with acquired analog channels to investigate timing errors or other signal integrity issues that are root cause of data errors.

Serial protocols that PicoScope can decode include 1-Wire, ARINC 429, BroadRReach (100BASE-T1), CAN, DALI, DCC, DMX512, Ethernet 10Base-T and 100Base-TX,

FlexRay, I²C, I²S, I3C, LIN, Manchester, MILSTD-1553, PS/2, SENT, SPI, UART (RS-232/ RS-422/RS-485), and USB protocol data as standard, with more protocols in development, and available in the future with free-of-charge software upgrades.

Data can be displayed in Hex, Binary, ASCII or Decimal formats. To help make decoded data even easier to read PicoScope enables use of a link file so that, for example, address hex 03DF can be displayed as “Oil Temperature,” or whatever the parameter means. A filter function displays only those packets that match userdefined condition(s). Statistics displays detail timing and voltage information on each packet, which helps to determine safe margins, noise immunity, and design reliability over extended operating periods.

References Pico Technology, www.picotech.com/

Real-time PC-based scope features: PicoScope 2000 Series: Smallest form factor enclosure that fits easily in a laptop bag and can be taken anywhere. USB powered and with bandwidths from 10 to 100 MHz, two channels or 2+16 channels MSO models. Ideal for education or for making waveform measurements on the move.

PicoScope 3000 Series: General purpose scopes to 200MHz with 2 or 4 channels plus 16 digital channels MSO models. A workhorse in the lab and perfect for home working.

PicoScope 4000 Series: High-precision scopes with fixed 12- and 16-bit resolution available. > 70dB+ dynamic range and differential eeworldonline.com

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input models available for making precision measurements from a few mV to hundreds of volts on non-ground referenced systems.

PicoScope 5000 Series: Versatile FlexRes scopes that can be set to make high speed timing measurements with 1ns resolution @ 8-bits vertical resolution or 12/14/15/16-bit vertical resolution at lower sampling rates. Suitable for laboratory or field use where there is a wide range of tasks to be performed day-by-day.

PicoScope 6000 Series: High-performance, deep memory scopes for demanding signal integrity and other scientific applications. Up to1GHz bandwidth and 4GS capture memory for performing big waveform data analysis.

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The many measurements of power quality There’s a confusing variety of ways to quantify power consumption.

Make sure you understand what your favorite power metric means.

Leland Teschler, Executive Editor

To

most engineers, the concept of power quality

pertains to characteristics of the ac line feeding grid energy to equipment in a facility. But variations of the “power quality” term now can refer not just to the ac line but also to properties of devices connected to the ac mains.

Thus it may be useful to review the basics of power quality and some of its newer interpretations.

First consider an “ideal” threephase power system. Here the current is in phase with the voltage for each phase, the phase voltage and currents are exactly 120° apart and all equal to each other, the voltage and current sine waves are not distorted the source impedance is zero, so events at the load don’t affect the source voltage, and the actual frequency is equal to the nominal frequency.

Of course, no real-world power system is ideal. There is an acceptable range of deviation.

In the U.S. ANSI C84.1 defines the acceptable limits for service voltage and utilization voltage.

Power Factor is one of the properties it defines. A power factor of one is ideal, meaning ac voltage and current sinewaves are exactly in phase with one another. Capacitive and inductive impedances tend to cause power factors less than one. Motors, solenoids and pumps typically have impedances containing inductive reactance, which vary with their mechanical load. Capacitors have

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impedances that are combinations of a typically small resistance and larger capacitive reactance component. Reactance present in ac systems shift the voltage and current sine waves out of phase from each other. Voltage leads current with inductive reactance and current leads voltage with capacitive reactance. Low power factor tends to arise in industrial facilities containing numerous motors or other inductive loads.

Electric utilities typically charge large industrial and commercial customers a higher rate for low power factor loads.

Unbalance arises in threephase power systems when single phase loads (lighting, office equipment, etc.) draw unequal amounts of current on each phase.

This kind of load causes greater stress on the neutral conductor.

Ideally loads are balanced, meaning the voltage and current phases are exactly 120° apart from each other, though the currents might not be in-phase with the voltages.

A balanced three-phase four-wire

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3DFS defines its PQR metric for each of the three phases and factors in both THD and phase lag/lead (cos £) between voltage and current. wye system will have zero current on the neutral wire. The amount of current on the neutral wire in an unbalanced system rises with the unbalance, potentially causing overheating and a risk of fire.

Motors driven by unbalanced voltages generate a phenomena known as counter-torque where a small motor torque works in the opposite direction from the motor rotation. Thus part of the energy delivered to the motor works against itself.

Harmonics are a form of waveform distortion arising in circuits containing non-linear loads primarily characterized by switching power supplies. These non-linear loads may impose higher frequency sine waves on the ac input, causing a power

lose in the form of wasted heat.

The excess heat produced by harmonics can be detrimental to a power system. Transformers are especially susceptible to damage caused by harmonic eddy currents which circulate in the iron core and produce excess heat.

Harmonics are multiples of the main frequency, 60 Hz in the U.S. For example, the third harmonic in a 60 Hz system is 180 Hz and the fifth harmonic is 300 Hz. Power quality meters can display the magnitude of each harmonic frequency. They may also read out total harmonic distortion (THD) and total demand distortion (TDD) to provide a single harmonic distortion measurement rather than an entire spectrum.

The point of the PQR measurement devised by 3DFS is to make real-time corrections in power consumption as in this example.

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POWE R ME ASUREMENTS PQR The widespread use of power quality metrics has given rise to more specialized but similar measurements often aimed at individual ac loads or ac-powered appliances. One such measurement is the power quality rating or PQR, a metric created by N.C.-based engineering firm 3DFS. An electrical network with a PQR of 28% utilizes 28% of the energy consumed as electricity and the remaining 72% as heat or vibration somewhere in the electrical network.

In a nutshell, 3DFS claims ordinary power quality measurements are incomplete because they typically are calculated by some combination of harmonic distortion and reactive power. The problem, says 3DFS, is that this measurement typically doesn’t consider whether there are imbalances across phases. PQR is meant to be a more informing metric that includes such conditions. The point of this exercise is to correct for harmonics and reactive power while simultaneouly balancing the phases as power flows in rel time. This is something 3DFS claims it can do via something it calls a software-defined power controller.

To complicate matters somewhat, there is a similar sounding metric called a power quality score, PQS. This one comes from Bijou Electronics in New York. Rather than being a metric for power consumption in a facility, PQS is aimed at individual power consuming appliances. Bijou sees PQS as potentially an energy efficiency benchmark for any device that gets plugged into the power grid. PQS is a combined metric containing power factor, efficiency or efficacy, and THD. PQS is a single

number rating of power quality at each device setting or at several load settings, typically 0, 0.1, 10, 25,50, 75, and 100% load for a power supply or power adapter. Bijou has published PQS results for a number of consumer devices and typically gives results at three points, idle, average, and max PQS. Bijou defines PQS as having a value from 0 to 200—it says the reason for going to 200 rather than something smaller is to provide more detail when comparing products. This helps find differences between various power consuming devices of similar types which may bunch up if made with the same basic topology.

Though any device can theoretically get PQS, Bijou is targeting smaller power-consuming devices, enforcing a 2.4 kW or 2.4 kVA upper limit. Bijou tests consumer devices and assigns a PQS score and publishes the results online. As of this writing, the firm has published results for products that include various light bulb technologies, fans, power adapters, transformers, humidifiers, power supply modules, display monitors, and even an electric blanket.

References Bijou Electronics, bijouelectronics.com 3DFS, 3DFS.com

CONNECT WITH US! CHECK US OUT on ISSUU.COM!

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Measuring ESR and ESL of dc-link capacitors

Tim Ashworth, Meng Li, Zurich Instruments

Impedance analyzers work better than LCR meters when it comes to sizing up the super-low impedance and inductance of modern power converter capacitors.

A dc-link capacitor used in a typical three-stage power inverter system on hybrid/electric vehicles. The dc-link capacitor is the key element in such a design to stabilize the dc voltage.

DC-link

capacitors are often found in power

conversion applications such as inverters, motor drives, medical power supply equipment, and so forth. Consequently, dc-link capacitors often need a high capacitance with a high dielectric strength. From an energysaving perspective, they should also exhibit a low dissipation factor (equivalently, a low equivalent series resistance, ESR).

More importantly, dc-link capacitors quickly stabilize the dc voltage. A low equivalent series inductance (ESL) is critical for this role. Thus it is important that design engineers understand ESR and ESL and how to accurately measure them in dc-link capacitors.

Consider a typical drive system on an electric vehicle. It includes a rechargeable bank

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of batteries, a three-phase inverter, and an electric traction motor. The battery bank can output a voltage as high as 800 Vdc and control the power delivery using power transistors (IGBTs or power MOSFETs) that switch in the kilohertz frequency range. A dc-link capacitor sits between the dc power source and the switching circuit. It is designed to provide a stable dc voltage by minimizing the voltage ripple that the inverter sporadically demands.

The electrical response of any “ideal” capacitor is purely capacitive with a phase angle of -90°. But a real-world device also has parasitic inductance and resistance. For an easy understanding, we can visualize an ESR and an ESL in series with the capacitor. The ESR is a lumped parameter representation of the dielectric loss, and the ESL corresponds to the inductance. ESL generally arises from the leads connected to the capacitor as well as from the construction (e.g., winding) of the capacitor itself. It is important to note that even 6 • 2022

An EPCOS TDK dc-link capacitor connected to the MFIA Impedance Analyzer using a custom low-ESL fixture.

The dc-link capacitor has three sets of electrodes labeled U, V, & W.

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L IN K CA PACI TO RS

Screenshot of LabOne showing two measurement sweeps of the short after running the fixture compensation routine. The frequency range was split into two parts: red trace from 1 kHz to 100 kHz in the upper sweep showing real(Z), and green trace from 100 kHz to 5 MHz showing series inductance. The result confirms the low baseline for both the real part impedance (ESR) and series inductance (ESL) for this measurement setup. with three equivalent elements, this model can sometimes be too simple. As a result, the ESR and the ESL may not stay constant over frequency.

The ESR and the ESL of a dclink capacitor (and its connectors) play a significant role in the aforementioned EV drive system.

The ESR causes power dissipation and generates heat, which can lead to overheating problems.

On the other hand, the ESL stores inductive energy. When the motor drive transistor switches off, the resulting transient can cause voltage overshoot.

Thanks to technology advances in manufacturing, dc-link capacitor ESL can be reduced to the nanohenry level and ESR to less than 1 mΩ. But designers will want to confirm the values on the spec sheet with accurate measurements to ensure the capacitor can meet safety requirements with enough margin. eeworldonline.com

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impedance at a fixed frequency, whereas an impedance analyzer-such as the Zurich Instruments MFIA Impedance Analyzer--can measure impedance as a function of frequency. An impedance analyzer is generally more desirable as neither the ESR nor the ESL of a dc-link capacitor are constant, and we want to study their frequency dependence.

The MFIA provides the LCR function while also being able to sweep frequency and graphically display the acquired impedance parameters. To measure the ESR and the ESL, we mainly use two tools provided in Zurich Instruments software called LabOne: the sweeper and the compensation advisor. The sweeper allows sweeping parameters of interest (such as frequency) in a freely adjustable number of steps. The

compensation advisor promotes accurate and reproducible measurement results via a step-by-step guide to eliminate parasitic impedance from the test fixture or cabling.

As an example, consider a device-under-test (DUT) capacitor from TDK having a nominal capacitance of 120 µF, an ESR of 0.8 mΩ, and an ESL of <15 nH. We might connect the DUT to the instrument’s front panel via a custom fixture. The fixture features flexible connectors to allow for the vertical offset of the dc-link busbar connectors which match the IGBT module design from the manufacturer. The other end of the fixture uses four BNC connectors in a standard 22-mm spacing.

The first step in accurately measuring the impedance of a dclink capacitor consists of running a compensation routine that allows

LCR meters and impedance analyzers Both LCR meters and impedance analyzers can characterize dclink capacitors. Both instruments measure impedance based on Ohm’s law by taking a ratio of phase-sensitive voltage to current.

The ratio gives the absolute impedance and phase from which the real and the imaginary impedance can be calculated.

Parameters such as capacitance, inductance, resistance, quality (Q) factor and its reciprocal (dissipation factor, DF) are subsequently derived from the complex impedance. They are derived by applying a proper equivalentcircuit model on the measured impedance. LCR meters and impedance analyzers often include several built-in models to extract these parameters.

Nevertheless, the two instruments are different: An LCR meter primarily measures

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Screenshot of LabOne showing a wide frequency sweep from 1 kHz to 5 MHz of the dc-link capacitor. The five traces are: capacitance (dark blue), real impedance Z (green), absolute impedance Z (red), series inductance (light blue) and phase (purple). The self-resonance frequency of this capacitor is 90.8 kHz. The green real Z trace is equivalent to ESR. Below 90.8 kHz, the self resonance frequency (SRF) of the capacitor, the capacitance can be seen as the dark blue trace. At 1 kHz, the capacitance is read as 121.999 µF, which is consistent with the specified value of 120 uF ±10%. Above the SRF, the ESL is plotted as the light-blue trace. Here three peaks are annotated by black arrows at 175.9 kHz, 284.2 kHz and 749.7 kHz. This panoramic view is helpful to understand the capacitor’s behavior at different frequencies for a more reliably prediction of its response in real-world applications.

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Screenshot of LabOne showing two sweeper windows each displaying fifteen traces, corresponding to five measurements of each set of three electrodes. The upper sweeper window displays the ESR from 1 kHz to 100 kHz and the lower sweeper window displays the ESL from 100 kHz to 5 MHz. The traces are color-coded to group the three sets of electrodes (U red, V green and W blue). for the impedance of the fixture in the actual measured data. In this example, we can run a load-short routine from 1 kHz to 5 MHz. This procedure allows us to redefine the measurement plane to be at the connectors of the fixture which directly touch the capacitor.

Next, we can again measure the short to get an idea of the measurement baseline. The resulting ESR and ESL of the short can be separately displayed in two sweeper windows. In this example, these sweeps confirm a low baseline of 15.7 µΩ and 1.7 pH for the ESR and the ESL, respectively. Such low values give us the confidence that we can, subsequently, reliably measure the capacitor.

We can then simply repeat the first step by opening two different sweeper windows to cover two ranges: 1 kHz to 100 kHz and 100 kHz to 5 MHz. As the dc-link capacitor in this example has three different sets of electrodes, each electrode set is measured sequentially and repeated five times to demonstrate the repeatability of the measurements. The test signal amplitude is set to 900 mV. Under default settings, a sweep of 200 frequency points lasts around 12 sec.

The upper window of the nearby screen shot shows a sweep of real(Z) corresponding to the ESR. In total, there are fifteen traces in the sweeper, color coded to the electrode set. The traces overlap to a high degree thanks to the excellent repeatability of the measurement even after disconnecting and reconnecting. The ESR measured using the electrodes W (blue traces) can be read from the black arrow showing 718 µΩ at 11.35 kHz. This nicely agrees and confirms the stated value of ESR at 0.8 mΩ. The yellow trace in the sweeper corresponds to a short measurement as our baseline.

In the nearby screen shot, the lower sweeper window shows the ESL from 100 kHz to 5 MHz. Traces are color-coded to match the three sets of electrodes, and traces overlap because measurements are repeatable to a high degree. A light-green trace corresponds to a short measurement as the baseline in ESL and is much lower than the three electrode sets.

Electrodes U and W are geometrically symmetric and show a similar behavior with three peaks at approximately 176 kHz, 283 kHz and 742 kHz. The center set of electrodes, U, in contrast, has only two peaks in ESL. The lowest peak value of ESL from the blue traces is 9.49 nH at 742 kHz. This is again in a nice agreement with the stated value of <15 nH. Thus the ESR and the ESL of a dc-link capacitor can be accurately and repeatably measured using an impedance analyzer. After carefully compensating for the parasitic impedance of the test fixture, the measurement plane can be reset from the instrument’s front panel to the DUT’s position.

In this case, measurement baselines of both ESR and ESL are significantly lowered, to 15.7 µΩ and 1.7 pH, respectively. This leaves a lot of headroom for future measurements as manufacturers continue to improve both ESR and/or ESL. In addition, the impedance analysis shows the variation of ESR and ESL as a function of frequency. The measurement not only confirms the specified values by the manufacturers, but also enables a detailed study of the DUT with circuit modeling.

References Zurich Instruments, www.zhinst.com/en

Getting an overview The next step is to measure the dc-link capacitor over the full frequency range of interest. The result is illustrated in the nearby screen shot with a multi-trace sweep from 1 kHz to 5 MHz. One trace shows the real part of the impedance and equivalent to ESR. At the lowest frequency of 1 kHz, the capacitance is read as 121.999 µF, which is consistent with the specified value of 120 uF ±10%. Above 90.8 kHz--the self resonance frequency (SRF) of the capacitor--the ESL is plotted and shows three peaks annotated at 175.9 kHz, 284.2 kHz and 749.7 kHz. In addition to the ESR and the ESL, traces show absolute impedance and phase.

This panoramic view is helpful to understand the capacitor’s behavior at different frequencies so we can more reliably predict its response in real-world applications.

Thus, we conclude that the ESR should be measured at low frequency, whereas the ESL should be measured at higher frequency.

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Automotive cable and harness testing made easy A hipot tester, switch, and automated software can assure vehicle connectors are safe and reliable.

Loose

or mismatched

Ground testing

electrical

Ground testing can employ either a continuity test or a ground-bond test. Both measure continuity and the resistance to ground.

However, with ground-bond testing a high current runs through the grounding system to ensure the capacity to ground properly without melting open. Depending on the source current (50 mA to 40 A), the user can test for:

components cause the highest rate of failures in vehicles. Often harnesses and components are inadvertently pinched, miswired or damaged during installation. Also, cable insulation that isn’t secured can wear through and cause eventual problems.

For these reasons, it is critical that automotive cable and harness assemblies be thoroughly tested and verified before installation. One proven approach to cable/harness testing combines a hipot tester, a switching system and automated software. This testing system speeds and simplifies testing in automotive applications, while also providing historical backup data to prove that components were verified prior to installation.

Vehicle harness testing verifies cable insulation for isolation, sufficiency and effectiveness. A simple insulation resistance test, also referred to the dc hipot test, employs two conductors with an insulator between them. One conductor can be a vehicle chassis, for instance, and the other a 12-V positive terminal coming off the car.

During isolation testing, high voltage is introduced to the limit desired. The test ramps voltage up from zero to 1,000 V, holds it for a specific time period, then lets it discharge. During the test, a hipot tester monitors the voltage to assure that excessive current does not flow from the conductor through the insulation to the other conductor.

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• •

• •

Type test or production Resistance from chassis to earth with either twowire or a four-wire Kelvin measurement Circuit breaker trips properly 100 mΩ typical limit

Typically in continuity groundbound testing, a four-wire Kelvin measurement test is used to measure low contact resistance.

During testing, the hipot sources current across the DUT, measures the voltage drop across it, and automatically converts the result into milliohms.

In contrast, for ground-bond testing, the user pushes high current through the cable. For example, the conductor may be stressed with current levels up to 40 A, potentially generating heat. A faulty connection will cause runaway and failure.

Performed during installation, this test is key for avoiding future vehicle performance problems and, potentially, unsafe conditions.

The combined solution described here comprises three types of Vitrek test equipment: a hipot tester, a multi-point switch,

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Glen Broderick , Vitrek Inc.

Isolation testing using hipot tester.

and automation software. In this example, the 95X Series hipot tester performs insulation resistance, continuity and ground bond testing.

This unit is well suited for measuring lower leakage current levels in higher-voltage harness tests.

The hipot tester offers a wide range of ac/dc outputs (up to 30 kV ac and 15 kV dc) and 100 pA resolution leakage measurements.

The system can be used with or without the software because the 95X Series is capable of controlling itself and the 964i switch as a standalone system.

(With the addition of a computer and QT Enterprise software, all test data can be archived for later reference or audits.)

Also part of the combination cable/harness test system is a multi-point, multi-conductor 964i switch. The switching system offers a range of configurable voltage ratings (3,7,10 and 15 kV) with currents up to 70 A. The unit has up to eight relays per card for a total of 64 test points. Multiple chassis can be combined to increase channel count beyond 64 points.

Each 964i is custom wired per user-specific test requirements.

For instance, some people like “Y” connections where each point can be made either HV or return.

Others like one-to-one inputs and outputs that can be turned on and off individually. Still others like to have an internal HV or return bus.

Example of four-wire measurement lowresistance testing.

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CABLE TES TI NG Vitrek’s combination cable/harness test solution includes a 950X Series hipot tester, a 964i switch (below) and QT Enterprise software (screen capture).

Any combination of points can be connected to the bus at any given time.

Integral to the combined solution, QT Enterprise software offers extremely helpful control capabilities. Used with the test equipment described above, this software enables the user to view a graphical representation of their test on a computer screen – and configure all test parameters, such as voltage and output, on easyto-read display screens.

In the event of a failure during testing, the user can set the test to stop to either re-test with a new device or continue the test process to identify any additional faults before reworking the existing device. When testing is complete, the user can save and document (and print) an unlimited number of tests on a central database for comparing and analyzing the test results of various users.

Cable manufacturers will find the software’s “global” settings especially helpful because they allow companies to personalize their testing processes. For example, a user can specify that all testing will start upon a digital I/O (DIO) input or at the click of the mouse.

When the switch is paired with a hipot tester, up to 16 chassis can be daisy-chained with hundreds of unique test points. This means the tester goes from having one high-voltage output and one return point – to having hundreds.

Typically, the user will connect one highvoltage and a return lead to run a test. The user can custom configure and display each switch according to the number and type of cards.

Unlike typical cable testers, the system allows each relay to be configured individually, which eeworldonline.com

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means easier testing of complicated devices.

Use of QT Enterprise software helps automate testing by giving users computercontrol of their tests. For instance, cable/ harness tests can be configured according to specific performance parameters, including: • • • • • • • •

Meanwhile, isolation must be high, in the 5 kV range. The EV connectors, each with multiple temperature-sensing pins, etc., require at least three to five conductors each, which add to the complexity of the test.

After specifying the relay sequencing using QT Enterprise software, the application utilizes the 952i hipot tester’s high-voltage isolation test and ability to measure resistance.

The 964i switch measures contact resistance in microohms, tested at 5 kV isolation.

With 20 or more harnesses, many interfacing with digital control devices, today’s vehicles have complex electronics systems. They need precise and accurate continuity and cable testing on every connection during the assembly process to assure safety and reliability. A combination of a hipot tester, multi-point switching system and automated software shortens test time, boosts production volume and improves quality.

Automated cable testing reduces the chances of error from manual testing. It also provides manufacturers with unlimited historical data that can serve as valuable backup in case of a subsequent component failure.

References Vitrek Inc., vitrek.com

Up to 15 kV Up to 70 A 64 relays per chassis Up to 16 chassis 100 msec/point continuity test 1 sec/point voltage-withstand test Each relay is individually controllable Can be ordered with configurations to connect any way each specific user requires

Example: EV charger testing The automated harness testing system is used in an electric vehicle charging cable application. The equipment/software used is a Vitrek 964i high voltage switching system (and HVHC relays), a Vitrek 952i hipot tester, and Vitrek QT Enterprise software, The testing validates EV charging cables and verifies ultralow contact/cable resistance and HV insulation.

For the test, the cable sat inside a safety enclosure to ensure no high voltage was applied while the lid was open. Although the hipot testers have built-in technologies to limit discharge and quickly detect breakdown voltage, the enclosure provides additional operator safety.

Because a tremendous amount of energy goes into the system as fast as possible, the connectors must have a low contact resistance.

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Vitrek’s QT Enterprise software provides a graphical representation of cable testing.

When testing cables, the user can individually control each relay. All 64 relays connected to the 964i switch are represented on the screen. If there is more than one chassis, each chassis can be selected individually and its outputs changed. When viewing custom configuration, the user can determine which relays should be shut and which should be open for each test. Here, the user has selected boxes “1” and “33” in the blue “Bank” columns. During testing, the user can view the steps comprising the testing sequence. After the test is complete, the user can easily choose the switch setup for the next test.

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Measuring power choke inductances Hubert Kreis, ed-k • Alan Lowne, Saelig Co. Inc. The inductance of power chokes changes with current level.

Conventional small-signal-measuring bridges are out of their depth here!

With

the exception of air-core coils, all power inductances show

a saturation behavior; their inductance drops with rising current. The core materials can lose permeability and then behave like an air core in extreme cases. That saturation behavior limit can be influenced by the choice of core material, the core geometry, the number of turns, and the air gap.

There are often deviations between the calculated inductance at a certain current (e.g. the nominal current) and the real inductance, perhaps because the choke geometry causes a non-homogeneous field distribution, or the data sheet information of the cores is incomplete.

Manufacturing variations of core tolerances are often noticeable, as well as temperature influences. So the saturation behavior of power chokes must be measured during product development and also during quality inspection.

Power chokes serve in many applications: smoothing chokes for switch-mode power supplies, filter chokes for IGBT converters (sine filter), an impedance for line-commutated converters, a smoothing reactance for dc circuit blocks, and so forth. A common application is as a smoothing choke in the secondary of a switchmode power supply. Here the voltage output from the switch typically has a rectangular shape. The output current is basically dc with a superimposed ripple that resembles a triangular wave having a frequency corresponding to the circuit clock frequency - from a few hundred hertz up to several megahertz.

For the circuit designer, the spec sheet value for the inductor in this circuit is usually not particularly useful. More meaningful is the inductance at the highest-occurring direct current. Inductance at this level affects the superimposed ripple current (and thus the residual ripple of the power supply) as well as the maximum current through the switching power semiconductor. If the choke effect goes into saturation before reaching the desired maximum output current, bad things happen.

Power semiconductors can be damaged or overheat, the output capacitor could overload, and the ripple on the output would rise sharply.

The problem is basically the same in other circuit topologies and applications employing power chokes, such as sinusoidal filters for IGBT converters. Standard small-signal-measuring bridges don’t provide insights into inductor behavior at high currents because they only measure the initial inductance using extremely small measurement currents. Measurement of the saturation behavior requires putting a corresponding high current through the choke.

It is also important to realize that the inductance of each choke is frequency-dependent.

Fixed frequency measurements There are basically two different measurement methods for inductance: the fixed-frequency method used by LCR meters and the pulse measurement di/dt method. Fixed-frequency methods feed a direct current to the DUT. A sinusoidal small-signal measurement voltage (e.g. 10 kHz) is superimposed on the current, and the inductance is calculated from the amplitude

and the phase of the measured current.

The advantage of this method is that the measurement frequency is set precisely and reproducibly. However, the problem here is that the measurement conditions have little to do with the real-world application conditions.

Power chokes don’t normally see a low-voltage sinusoid but rather a rectangular voltage containing multiple harmonics. Real-world

Typical application of a filter choke in a switch-mode power supply. conditions require a powerful dc source, which can be expensive for larger currents (e.g. over 20 A). Moreover, inductor characterization over the whole direct-current range requires many individual measurements taken at different current levels.

In contrast with fixed-frequency methods, measurements with the pulse di/dt method subject the test object to a square-wave voltage pulse mimicking that in the real-world.

Current is applied to the DUT while both the current di/dt rise and the saturation behavior is observed for the inductance estimation.

The measurement pulse stops when a preset maximum current is reached. Evaluating the rate-of-rise di/dt of the current creates a complete inductance curve for the test object with a single pulse.

Ideally, the pulse voltage corresponds to what the device sees in the real-world to

The ed-k DP10 Power Choke Tester.

Typical voltage and current curves for a filter choke.

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IN DU C TO R TES TI NG the inductance curve) relies on equation one.

Increments for Δi or Δt must be carefully chosen. If the step size is too small, even small digitization errors will result in fluctuating inductance curves. Step sizes that are too big won’t accurately reproduce sudden saturation phenomena. What is needed, therefore, is a dynamic step size control in which the step size automatically adapts depending on di/dt results.

Fixed Frequency Method measurement setup: The test object sits in a measurement bridge and is biased from a direct current source. C∞ isolates the component from the measurement bridge. avoid the error-prone results from a small-signal fixed frequency measurement. An additional advantage of the pulse-shape measurement is that the current source need not deliver the test current continuously; a capacitor bank can deliver the test energy. This approach saves considerable cost and reduces the size of the test instrument.

A point to note is that the voltage of the measurement pulse at the DUT is never constant because of parasitic voltage drops on the supply lines. The ohmic resistance, RL, of the DUT itself must also be taken into account though the DUT parasitic capacitance CL can almost always be neglected. The following equation then gives inductor inductance LL: LL(i) = [UDUT (i) - RL × i] × dt/di (1) The pulse method employs individual measurement pulses. Thus, the voltage at-and the current through--the DUT both must be recorded for evaluating the shape of the inductance curve. This recording takes place via a fast and accurate A/D converter with high resolution.

In plots of measurement current and voltage using this method, it is often possible to at first glance see where a choke goes into hard saturation based on the slew rate of the measuring current. For a more precise result, or the creation of an inductance curve, a numerical evaluation necessary. The signal evaluation (the calculation of

Measurement setup with the pulse measurement di/dt procedure. The output current is not galvanically connected. eeworldonline.com

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Measurements in practice Its inherent advantages have made the Pulse Measurement procedure a standard for assessing the true inductance of power chokes worldwide.

The DPG10 Power Choke Tester from German company ed-k embodies the pulse measurement method for currents from 0.1 to 4,000 A, allowing measurement of the saturation behavior for even large power inductances. The voltage of the measurement pulse can be set from 10 to 400 V, so any inductance can see a voltage present in

Measured current and voltage curves on a choke with an amorphous tape-wounded core and air gap (Lrated = 190 μH, Irated = 135 Arms) for active power factor correction . The choke goes into saturation at around 200-250 A. Here the voltage of the measuring pulse is around 100 V. When the measurement current reaches 500 A the pulse terminates. Based on the slew rate of the measuring current it can be seen at first glance that this choke goes into hard saturation at around 200 to 250 A. the real peak current after the superimposed current ripple is figured in. For the circuit designer this underspecification can have negative effects on the power semiconductors, the line perturbations, and the losses.

Extended equivalent circuit of the measuring arrangement for di/dt procedure. the real application (e.g. filter inductor for a sine inverter output around 400 V, or a smoothing choke for a 5-V output of an ac/dc converter around 20 V). The duration of the measurement pulse can be pre-set from 3 µsec to 70 msec.

The maximum possible pulse energy is limited by the internal capacitor bank supplying the measurement current. At maximum measuring voltage, it is up to 8 kJ. This is enough even for large power chokes.

The DPG10 Power Choke Tester with its fast IGBT switch design handles almost all types of inductive components, from small SMD chokes to power chokes in the MVA range weighing several tons. An attached PC allows the DPG10 to make automatic inductance measurements, displaying the results that include ohmic resistance. The results are available quickly, and test procedures don’t heat up the test specimen.

The measurement protocol determines the inductance curve as a function of current, both as a diagram and in tabular form. It is not unusual to see a nominal value of inductance that is well below that of the inductance at

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References ed-k, www.ed-k.de/en Saelig Co. Inc., www.saelig.com

Saturation-dependent inductance curve of the same PFC choke. This choke at a peak nominal current of 135 A × 1.41 = 190 A has an inductance of 156 μH. However, the real peak current in the application is about 30 A higher due to the superimposed current ripple. The inductance at 220 A is then only 127 μH, 33% below the specified nominal value of 190 μH.

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Debugging digital and analog signals with cross-domain analysis Fernando Gomez, Tektronix Inc.

Multi-domain signal analysis is blurring the line between real-time oscilloscopes and real-time spectrum and signal analyzers.

Multi-domain

signal analysis is enabled by advanced oscilloscope features such as 12-bit ADC, digital down conversion (DDC), and flexible channel inputs where each can be either analog or digital. As Signal Integrity (SI) and Power Integrity (PI) engineers know, real-time oscilloscopes will provide frequency domain information by executing a Fast Fourier Transform (FFT) having accuracy and resolution that are direct functions of the sample size/ record length. Unfortunately, as the FFT is a post-acquisition operation, there is no practical way to understand the shape of the frequency response relative to the time span during which the analog signal was acquired.

At this point, a logical next step might be to use a spectrum analyzer (SA) to provide a direct capture of the signal under test. For many years, traditional swept SAs have provided radio frequency (RF) information for steady-state, time-invariant signals. But vector signal analyzers (VSA) can also capture and record signals over time, helping to examine RF behavior including modulation analysis or pulse analysis over the acquired time span.

Finally, real-time spectrum analyzers (RTSAs) add real-time performance and include VSA analysis plus trigger capabilities as well as time-resolved analysis of amplitude, frequency and phase over time.

These frequency domain instruments

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can provide the information needed.

However – significant limitations must be acknowledged: Acquisition bandwidth is limited mostly to 1 GHz, and these instruments are limited to a single channel so they cannot look at multiple RF signals simultaneously or correlate to other signals in the system Of course, real-time oscilloscopes can address these limitations with wideband capture, multi-channel operation and, in some cases, provide time-correlated multi-domain operation. So should scopes be used for RF work?

There are a few caveats worth considering before embarking on RF measurements using oscilloscopes.

Is there enough dynamic range to capture the signal’s full energy content?

Oscilloscopes are typically limited by the bit depth/vertical resolution of the ADC. RF signal analyzers normally feature 12-16 bits ADC resolution, at narrow bandwidth captures – up to 1 GHz. When wider bandwidth is required, however, the dynamic range is compromised. Thus there must be a tradeoff between high vertical resolution and bandwidth coverage.

How deep is the channel memory?

In a VSA or RTSA, memory is efficiently used because the RF signal is sampled and down converted, resulting in fewer samples stored while preserving the frequency content of the signal.

An oscilloscope, on the other hand, uses direct sampling of the RF signal.

It can oversample at multiples of the fundamental frequency to ensure adequate frequency coverage, but the consequence of storing tightly spaced 6 • 2022

FFT and Spectrum View Displays

Time Domain view and resulting FFT

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C R OSS-DOM AIN A NA LYS I S

Digital Down Converter (DDC) Block Diagram

sample points is that the memory fills out quickly. So only short time spans of the signal can be captured.

What about preselection?

RTSAs and VSAs usually incorporate preselection, or the ability to filter out second and higher order images from the acquired spectrum, at their front ends (FEs).

However oscilloscopes don’t normally feature this capability, and preselection is not generally supported at higher frequencies in either class of instruments.

We will consider a simple RF signal measurement using a real-time oscilloscope. Because oscilloscopes capture and display waveforms by default in the time domain, we will make use of an FFT (usually available as a math function) to obtain the frequency domain representation of the signal. In this case, we will be acquiring a clock with a nominal frequency of 98 MHz. First we display a few cycles of

this clock signal in the time domain and compare them with the FFT display with the same center frequency and span. Here, the FFT provides little to no information because the resolution of the FFT is inversely proportional to the length and time of the vector.

This means the only way to realize higher resolution is by acquiring a higher number of samples. Doing so provides a spectrum that looks good, but the time domain view is not particularly useful. Actually the time domain view and the frequency domain views cannot be optimized simultaneously!

How do we resolve this limitation? Let's consider a digital down converter (DDC). The DDC numerically mixes the ADC samples down to baseband-- in other words, it removes the carrier from the signal, leaving the signal containing the information around dc, or baseband. The baseband data is formatted as I (in-phase) and Q (quadrature) component data.

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As data comes from the ADC, the numerical clock oscillators “project” the signal into two parallel processing paths. I and Q data points are represented in the phasor diagram, where the length of the vector is its magnitude and the angle is its phase.

By storing I and Q in memory, we can find out what the spectral content of the signal (magnitude and phase) is at any point in time.

Moreover, this I and Q data is recorded into memory in parallel with the ADC (time-domain) data, and the storage requirements are much smaller compared with the ADC data.

At this point, the DDC

independently of the ADCsampled data. This is available in every channel, with the same or different center frequencies.

To summarize, incorporation of the DDC in the acquisition ASIC resolves the issue of optimizing the view of time and frequency domain signals. It also allows for a simultaneous, time-correlated time and frequency domain capture.

Moreover, it makes possible independent acquisition settings and controls in each domain So what does multi-domain, time-correlated signal analysis mean at the system level? The answer is a full, 360° signal analysis via any combination of analog and

PLL Block Diagram

Multi-Domain Signal Analysis: PLL Example data is time-correlated to the ADC data. The DDC only needs a sample rate sufficient to cover the bandwidth of the information signal (IF). In conclusion, by “basebanding” or downconverting the signal we obtain the benefit of larger, deeper storage of RF content with significant lower memory requirements – as DDC data samples will be much further apart in time.

Suppose we now go one step further and incorporate the numerical DDC into the acquisition hardware. Then I and Q data, along with the ADC data, is acquired in every channel and becomes available to the acquisition memory and the trigger system. By placing the DDC and ADC data into memory, we now have the two data sets correlated in time.

Additionally, the spectrum processing vector (time required to capture the frequency domain signal) can be managed 6 • 2022

digital signals, parallel and serial buses, math waveforms, and RF signals and triggers. All of these signal inputs can be analyzed and measured in a time-correlated way with independent adjustments to optimize their analysis and viewing.

It may be useful to review how the RF capture and multi-domain analysis capability can be useful for digital designers. The benefits may not be immediately obvious.

But consider the example of embedded systems with phaselocked loop (PLL) circuitry. PLLs are commonplace in today’s mixed-signal designs. PLLs are closed-loop feedback systems consisting of both analog and digital components including a voltage-controlled oscillator.

In a nutshell, PLLs are used for the generation of an output signal, the frequency of which is synchronized (locked) to that of a reference input. As different clock frequencies can be obtained with these control loops, PLLs are

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frequencies can be obtained with these control loops, PLLs are increasingly used in microcontrollers to manipulate and control the frequency of clock signals.

The basic PLL circuit involves a phase detector at the input, a low-pass filter (LPF) and a voltage controlled oscillator (VCO) at the output, which feeds an error signal to the phase detector, thus completing the loop. Suppose we capture output signal with an oscilloscope having dual ADC and DDC signal paths. One can now investigate the relationship between the analog VCO tune voltage and the transient changes in frequency of the time domain signal. And this analysis can take place in sync with the I and Q sampled data (actually demodulated, or down-converted to baseband), and also at any location over the entire time domain waveform acquisition.

Information across signal buses, time domain and RF traces can be analyzed in sync, and the behavior of the control signals can be validated to determine correct operation.

Another example helps clarify how multi-domain signal analysis can benefit RF designers, who may be most familiar with using spectrum and vector signal analyzers. Consider a case where there’s a need for simultaneous phase or time-correlated RF signal capture across multiple channels. At the system level, signal sources could be multiple sensors or multiple antennas, electronic beam steering, multiple input multiple output (MIMO) applications, radar, electronic attack (EA) or electronic warfare (EW).

At the component level, multiple channels can be configured at the same frequency or at a different center frequency. For example, mixers and up/down frequency converters, harmonic distortion, intermodulation products, time-correlated spurious RF and LO feed-through characterization are just a few of many possible applications.

In a nutshell, Tektronix innovations in Real-Time Oscilloscopes allow for mix-and-match combinations of signal sources for the signal input with the FlexChannel system. This system enables simultaneous time and frequency domain signal capture. It does so by introducing a DDC in the signal acquisition ASIC on every channel and with independent control of the time and frequency domain settings.

As design complexities incorporate analog, digital and RF signals, multi-domain analysis is the perfect fit for design/ validation engineers with analog/time domain and RF/frequency domain backgrounds alike.

Multi-Domain analysis of RF signals: Mixer RF and time-domain outputs

References Tektronix Inc., www.tek.com

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5G test equipment goes mainstream Performance qualities once only available in exotic instruments can now be found in gear designed for production testing.

Rohde&Schwarz GmbH & Co.

5G

has now made the jump from research to mainstream adoption, and most chip manufacturers are well down the road to

developing second or third-generation components. In the early R&D phase, it was important to push one or two devices through millions of test scenarios. Early testing focuses on conducting every conceivable measurement to understanding what’s possible with the new 5G device.

But that changes by the time a part goes into production, when the testing regimine focuses on having a well-established set of parameters for what constitutes a validated part.

Today, most 5G device manufacturers are testing more products but running fewer tests on each one. R&D teams still need custom, lab-oriented equipment. But many RF engineering teams are at a point in the design cycle where they don’t need a full complement of test functions. Instead they need test equipment, specifically signal generators and signal analyzers, that can handle production-level characterization and validation at a lower price point.

In the R&D phase, 5G test and measurement equipment fulfilled a wide range of demanding requirements for component characterization in all specified frequency ranges. Tasks included supporting both sub-6 GHz and mmWave signals, while providing dedicated 5G NR measurement options.

These options fully supported the 5G NR flexible numerology, e.g. in terms of subcarrier spacing and multiple bandwidth parts.

Features included outstanding RF performance like high frequency and high bandwidth signal generation and analysis with extreme performing power flatness, phase noise reduction and EVM (error vector magnitude, sometimes also called relative constellation error, RCE) performance. Other aspects are full range of massive MIMO testing, ranging from true multiport real-time analysis of conducted cross-coupling effects at antenna arrays, going up to near-field and far-field over-the-air measurements.

R&D test & measurement systems that offer all of these capabilities are more sophisticated (and therefore more expensive) than equipment for 5G device production.

Production-level 5G testing As design margins shrink over the course of a design lifecycle, cost reduction can’t come at the expense of performance. In particular, accurate EVM is important because it represents some of the device’s most critical performance qualities. As a result, EVM performance has become a valuable benchmark for device evaluation. The benefits of getting EVM right cascades throughout the entire system.

Consequently, it may be useful to review a few EVM basics and how they relate to 5G system performance. An error vector, of course, is a vector in the I-Q plane between the ideal constellation point and the point the receiver decodes. Said another way, it is the difference between actual received and ideal symbols. Imperfections such as carrier leakage, low image rejection ratio, phase noise, and so forth deviate the actual constellation points from the ideal locations. The EVM is the root mean square (RMS) average amplitude

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Visible in the screenshot is an EVM of 1.22% (or –38.3 dB) and an EVM of 1.87% (or –34.6 dB) measured on the exactly same data. In the first measurement, the EVM is normalized to the peak power of the 64QAM constellation. The second EVM is for an RMS normalized measurement. The ratio of the values corresponds to exactly 3.7 dB. Below, a 64QAM constellation highlighting the difference between RMS and max power.

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5G ME ASUREMENTS of the error vector, normalized to ideal signal amplitude reference.

EVM =

Perror Preference

• 100 %

EVM is generally expressed either in percent or in dB. The ideal signal amplitude reference can either be the maximum ideal signal amplitude of the constellation or the RMS average amplitude of all possible ideal signal amplitude values in the constellation.

Unlike MER (modulation error ratio) for example, which per definition is normalized to the mean power of the reference signal, EVM normalization is not predefined.

While there is generally no confusion about the error power, there are two widely used versions of the reference power that can make a significant difference in the EVM reading.

In many cases, the EVM references the mean (RMS) power of the reference (ideal) signal. Some applications also use the peak power of the reference signal as the reference power. Obviously, there is no right or wrong here. It is more a question of the measurement task and the expected results.

Most generic Rohde&Schwarz measurement personalities give users a choice of reference powers. The (often unchanged) default setting is the RMS power. But a point to note is that in looking at only the symbol instants of a QPSK signal (where EVM is typically evaluated), there is no difference between RMS and peak power, because all symbols have the same amplitude.

For a 64QAM signal, use of RMS or peak power can make a significant difference – up to 3.7 dB. APSK or higher-order QAM modulations may result in even greater differences. The decision to use peak or RMS normalization depends on the application – but comparisons should only compare apples to apples.

Measurement challenges The higher frequency ranges of up to 54 GHz for 5G represent a steep jump in complexity

for testing and measurement. One of the inherent challenges with 5G is the highbandwidth signals that hit every element of the RF front end - antenna, amplifier, transceiver, etc. Measurement of integrated power over a particular bandwidth requires optimizing the signal-to-noise ratio by not over-ranging the instrument. When both the signal and noise are broadband, it takes instrumentation withi a high EVM performance over a wide bandwidth to make accurate measurements.

For example, the third-order intercept specification on a spectrum analyzer dictates the high end, which might not typically be a consideration for EVM. Because 5G signals are wideband with high peak-to-average ratios, it’s easy to exceed the power limit of the analyzer front end and get an error.

Test equipment with wide integrated analysis bandwidth that can sustain high RF performance-- including displayed average noise level (DANL) to third-order intercept point (TOI)--can significantly help. DANL, or sensitivity, as well as adjacent channel leakage ratio (ACLR) and spectrum emission mask (SEM) all serve as building blocks toward good EVM performance.

Insight into the dynamic range limits of RF components comes out of the abiltiy to see the lowest level signals in the presence of higher power, high data rate, and wide bandwidth signals. SEM performance is important because it identifies spurious signals, or spurs, that can originate either from both the device under test (DUT) and from the test instrumentation itself.

A spur that coincides with the frequency of the measured signal can produce inaccurate ACLR or EVM measurements.

Spurious signals become more frequent as you move downmarket from high-end test equipment to instruments built for broad adoption. But there are options available that deliver good spur performance at a midrange price. Phase noise is another important specification to consider in an RF test platform because it’s one of the main factors that can determine EVM performance. Good phase noise doesn’t guarantee good EVM, but poor phase noise will always result in poor EVM that cannot compensate for it in other areas.

To field mid-range instruments that Test equipment manufacturers are ushering in a new era of performance and cost-effective 5G high frequency testing with instruments like the SMM100A signal generator and FSVA3000 signal and spectrum analyzer from Rohde & Schwarz.

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perform well, test equipment providers are starting with high-end equipment that already provides excellent measurement quality and working backwards. This approach allows making decisions about which performance areas to scale back for reasons of cost. The resulting instruments still provide upper-range performance for the measurements that matter.

Starting from a high-end design does much more to preserve the quality of the measurement capabilities than starting from scratch and striving for some theoretical benchmark.

The latest signal generators and signal analyzers, thanks to this approach, represent a significant improvement over what has been possible even with high-end platforms.

It’s not uncommon to see instruments handling frequency ranges of up to 44 GHz and analysis bandwidth of 1 GHz, both necessary for the complexity of emerging 5G standards (3GPP Release 15 and beyond).

In fact, some of these specifications and techniques to optimize data rates require up to 1 GHz bandwidth. This requirement in itself represents a significant inflection point that will pressure test groups to upgrade their test systems to handle the latest standards.

A fast, integrated, out-of-the-box solution can help test teams cover a lot of ground and reduce the time to first measurement.

Multithreaded processing and server-based testing--where results can be processed centrally while the instrument is used only to capture IQ data--can help minimize test times.

In addition, automation features such as context-sensitive help and a built-in SCPI (standard commands for programmable instruments) macro recorder with a code generator helps speed production testing. As with many RF measurements, 5G production test instruments must also correct for the test set-up impedances. In this regard, it is helpful to use instrumnets providing an S2P file (s-parameter file) for all of the measurement components between the instrumentation and the DUT. This shift of the measurement plane closer to the DUT produces a more accurate measurement.

As 5G R&D shifts to 5G device manufacturing, test costs and speed become driving factors. However, performance still matters, so it’s wise to place a premium on instruments that can meet the demanding analysis bandwidth and frequency range coverage of 5G.

References Rohde&Schwarz GmbH & Co., www.rohde-schwarz.com

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Audio measurements for product development It pays to know the fundamental parameters that quantify the performance of sound equipment.

Quantitative,

objective audio measurements have a role to play not just during design validation and as a manufacturing quality control tool, but during all stages of product development . They even come in handy during the earliest stages of development, when a potential product is just a concept.

To understand why, we catalog the most common audio measurements made on complete audio systems and components and what they tell us: Level, Amplitude – Audio is generally understood to be an alternating signal, ac, in the band between 20 Hz and 20 kHz. When describing the amplitude of an audio signal, the general assumption is that we are discussing the root-mean-square (RMS) amplitude of a signal. That is, in the absence of any other notation, an audio signal described as having a level of 1 V can be assumed to be 1 Vrms. Because of this ambiguity it is helpful when describing the level or amplitude of a signal to note whether it is RMS - Vrms, dc - Vdc, Peak - Vp, or peakto-peak - Vpp as appropriate.

Frequency Response – An ideal audio device linearly reproduces the signals applied to it. There are many ways to make this measurement but the most common is to stimulate the device with a sine wave at one frequency, measure the level of the signal at the output of the device, and then move the test frequency. This is the classic stepped frequency sweep. In the modern era it is still the most common way to make this measurement. But it can also be made with a continuous frequency sweep, a so-called chirp. We can as well make this test with a continuous signal that combines

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Daniel Knighten, Audio Precision

many sine waves at different frequencies simultaneously. This can be a multitone, or it can be calculated using noise, music, or speech signals using a Fourier transform-based transfer function (FFT) calculation.

Regardless of how the measurement takes place, the result is a graph which displays the amplitude response of the test object as a function of frequency. If you had to assess the qualities of an audio device based on a single measurement, that measurement would be frequency response. The frequency response reveals whether a device sounds hollow or boomy, whether speech can be clearly understood, and most importantly is the first order indicator of product quality.

It is also worth noting that most of the additional measurements described here can take place at the same time as the frequency sweep used to yield the frequency response.

Distortion, THD or THD+N – No audio reproduction device is perfectly linear. The amount of energy in the output of the device, which is not linearly related to the input signal, is called distortion and is most calculated as Total Harmonic Distortion (THD) or Total

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Harmonic Distortion & Noise (THD+N). For both measurements, the device is stimulated with a pure sine wave at a single frequency.

The output signal of the device then has the stimulus signal or fundamental removed with a high-Q, high-attenuation notch filter.

The amplitude of remaining harmonic distortion products, THD, or the harmonic distortion products and noise, THD+N, are then measured. Commonly, distortion is expressed as a ratio to the total output signal level in percent or dB (decibels), but it is also sometimes expressed as the absolute amplitude of the component harmonics or total sum of the harmonics and noise.

For speakers, it is common to only report the THD. Most speakers are passive devices and have no sources of noise. A measurement of their THD+N will include unrelated background acoustic noise in the test environment.

For active electronic devices, it is common to report THD+N. All active electronic devices have sources of intrinsic A pure sine wave with a peak amplitude of 1.4 Vp and an RMS amplitude of 1 Vrms.

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AU DIO ME ASUREMENTS

Frequency response of a small bookshelf style speaker which demonstrates a good reproduction of sound above 200 Hz.

noise. And for some devices, the sources of internal noise may be greater than harmonic distortion.

When measuring distortion, it is critical to be unambiguous about what is being measured. In addition to clarifying whether it is THD or THD+N, it is important

At 1 kHz a single sample delay between two channels at 48 kHz sample equals 7.5° phase shift. to also include the test conditions such as stimulus signal level, output level, and the bandwidth of the harmonic products and noise included in the measurement.

Including or excluding harmonics and the bandwidth of the noise can make a radical difference in the measured result. This is one area which often makes it impossible to compare specifications from two different manufacturers. A specification stated as >0.001% distortion leaves too many questions open to be a meaningful gauge of device performance.

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audio quality metric beyond frequency response. Given two devices with identical frequency response, most listeners will identify the device with lower distortion as better.

Phase, Delay, Group Delay, and Polarity – Strictly, phase

represents the time alignment between two signals typically expressed in degrees of the period of those signals. Phase can be represented as the alignment between two or more output signals, as interchannel phase; or it can be taken from the input of a device to the output; finally, it can be absolute phase.

Uncontrolled offsets in phase between output channels can be extremely upsetting to listeners.

Humans use differences in phase or delay between our ears to locate objects in space. Listening to a familiar piece of music that has one channel slightly delayed compared to the other can be an amusing way to upset listeners.

A not-uncommon bug in digital signal processing software is to have

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one channel delayed from the other channel by a single audio sample. A phase measurement will easily detect this.

Absolute phase reveals the transition from capacitance to inductance in electronic components and the movement through the resonant frequency of speakers and microphones.

Group delay is the absolute phase of a device expressed as delay in time from input to output. Group delay reveals the effects of high-pass filters, especially ac coupling filters in electronics, and low-pass filters.

Group delay is a benchmark measurement in analog-to-digital and digital-to-analog converters.

Noise, Signal to Noise Ratio, and Dynamic Range – All devices that contain active electronics

have some degree of intrinsic noise. This intrinsic noise is a floor that sets the lowest level signal devices can reproduce. If you have ever heard white noise or powerline hum from a speaker when no music plays, you understand the issue. There are several different

ways to express this noise.

Noise, absolute or intrinsic, is the simplest. With no signal reproduced, any energy at the output of the audio device is captured, and the level of that energy in a defined measurement bandwidth is revealed.

When not playing a signal, some devices will mute themselves or go into a power-saving mode. It is also sometimes desirable to express the self-noise of a device relative to the maximum signal level it can reproduce.

Signal-to-noise ratio (SNR) and dynamic range are both ways of expressing this relationship. A classic SNR measurement first measures the maximum output level of a device, then the selfnoise. The SNR is then the ratio of the two. In a dynamic range measurement, the process is almost identical except instead of simply turning off the stimulus signal, a signal that is -60 dB to the fundamental is applied and then filtered from the output of the device under test. This ensures that the device does not mute itself when no signal is active.

Crosstalk – If an audio system has multiple channels, the degree of isolation between those channels is a standard measurement. The two-channel stereo system is the classic example of a multichannel audio system. Crosstalk is typically measured by stimulating one

Group delay of typical a/d converter displaying the effects of ac coupling filter.

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Power spectrum of a highperformance a/d converter displaying components of a THD+N measurement.

channel and measuring how much energy leaks into adjacent channels. High-crosstalk destroys positional information conveyed by multichannel audio systems; it collapses stereo imaging to mono-aural.

Sensitivity & Scaling – This is a significant figure of merit for any device that converts audio signals from one domain to another.

For a/d and d/a onverters, this scaling is the Volts-to-Full-Scale (V/FS) ratio between an analog signal and the digital audio sample values that correspond to it.

For microphones sensitivity is typically expressed as the Volts-output per Pascal of sound pressure (V/Pa).

Speaker sensitivity is usually described as a sound pressure measured at a certain distance for a given power input to the transducer, e.g., 94 dBSPL at 1 m and 1 W. However, in practice a power meter is seldom used. It is assumed that the speaker has a nominal impedance, and a specific voltage is used instead, e.g., 2.83 Vrms is applied to the speaker terminals for a nominally 8-Ω driver.

Directivity – When a single frequency response curve is all that is provided for acoustic devices, speakers, and microphones, it is safe to assume that the response curve represents an “on axis” value. That is, the measurement took place with the measurement microphone

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pointed directly at the device. Few speakers and microphones have a perfect, omni-directional response.

In fact, typically it is desirable for a device to project (speakers) or pick-up (microphones) more in some directions over others. Directivity plots are therefore the product of making many frequency response measurements of the speaker and microphone at different orientations.

This information is most commonly expressed in a polar plot which renders the response at different frequencies and different angles relative to the test article.

Impedance & Thiele-Small parameters – For loudspeaker drivers and systems, it is important to understand the impedance of the device and specifically the impedance as a function of frequency. It is common to hear speakers described as exhibiting a specific number of Ohms (frequently 8 Ω), but all real-world electro-dynamic speaker drivers have a complex impedance curve.

The impedance curve is important for determining two core requirements. First, what are the amplifier properties needed to successfully drive the speaker? A low-impedance driver will require a higher current supply while a higher-impedance driver will require higher voltage rails. Second, the impedance function is key to deriving the Thiele-Small parameters of the driver. These parameters

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establish the requirements for the mechanical enclosure that will yield the desired frequency response from the finished system.

Maximum Output, Overload Points – What is the highest signal level a device can reproduce?

For some devices the answer is relatively straightforward and perhaps even self-descriptive. If a DAC has a scaling value of 2.5 V/FS, then 2.5 V is the maximum signal it will reproduce. But what about analog amplifiers, speakers, or microphones?

For amplifiers, the traditional definition of the maximum output level is the largest signal amplitude the amplifier can drive into a load of X ohms with less than or equal to a certain distortion. E.G., 12 Vrms into 8 Ω, ≤ 1% THD+N. This figure may be modified by thermal limitations as it is also common to see max output defined in terms of a sine burst waveform. This figure reflects a design that can deliver a certain amount of peak power but only for some number of milliseconds.

For microphones, max level is typically defined as an overload point and often specified in absolute terms, e.g., Overload Point, 120 dBSPL. For microphones,

exceeding the overload point may permanently damage the diaphragm, so there are often no other conditions applied to the specification. The one exception is usually just distortion. E.G., 120 dBSPL, ≤ 10% THD. A variety of factors may go into determining the maximum signal level that a speaker can reproduce. Speakers may be limited by the mechanical travel of the cone, but also by the ability to dissipate heat during extended operation. Consequently, the largest signal a speaker can reproduce is often determined empirically. One technique starts by incrementing the signal level to a speaker and noting the point where higher signal amplitude does not produce a corresponding increase in output level compression. Another approach is to determine the maximum signal a device can reproduce for a given period of time without failure from thermal stress.

Using measurement data Measurements are the objective expression of subjective product requirements. In the earliest phases of product development,

Polar response plot of a typical loudspeaker.

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AU DIO ME ASUREMENTS it is useful to have a target frequency and distortion response.

Objective quantification of subjective criteria is especially important when undertaking the design of an acoustic device. For speakers it is generally, but not universally, accepted that when measured on-axis in an anechoic environment a flat frequency response is desired. For earphones and headphones, there is no universally accepted, desirable frequency response function. The human ear presents a complex acoustic load that is unique to every individual.

For earphones and headphones, it is especially important to create an objective target from the evaluation of subjective criteria.

Beyond the conceptual phase of product development, it is typical to begin qualifying components. At this point, there are two powerful imperatives for audio measurements. While component suppliers generally specify many parameters for parts they supply, those properties may not be paramount for your design. Perhaps more important, almost no two component suppliers specify parameters the same way.

During component qualification, measurements ensure apples-toapples comparisons among different devices. Measurements also help confirm components meet the specific requirements of your product.

In a nutshell, the chain of audio measurements support objective requirements that ensure a new product performs as intended.

Typical impedance curve for a small 2.5-in speaker and corresponding Thiele-Small parameters.

References Audio Precision, www.ap.com/

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Exceeding 100 Gbps with a sub-terahertz 6G testbed

Here’s a look at testing necessary for super-high throughputs involved in the emerging area of 6G. Greg Jue, Keysight Technologies

The sub-THz testbed for H-Band, 220-330 GHz. The testbed uses a 92 GSa/sec M8196A arbitrary waveform generator (AWG).

6G

research is in its early stages. The vision for what the International

Telecommunication Union calls Network 2030 continues to take shape. While the industry is years away from starting the standards development process, subterahertz (sub-THz) territory is a focus of active research.

Data throughputs of 100 gigabits per second (Gbps) to 1 terabit per second (Tbps) are key objectives for 6G. This extreme data throughput could evolve into a key performance indicator (KPI) for 6G. However, the throughput jump poses significant challenges for both RF and baseband technologies.

There are three fundamental approaches to increasing data throughput. The first employs higher-order modulation schemes such as 64 QAM to boost the number of bits transmitted for each symbol. Given a fixed and finite spectrum bandwidth, increasing the modulation order from QPSK (transmitting two bits for each symbol) to 64 QAM (transmitting six bits for each symbol), would raise the data

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throughput by a factor of three, if channel conditions and radio performance allow. A 1-GHz QPSK symbol rate would result in a 2 Gbps theoretical raw calculated data throughput without forward error correction (FEC) coding rate redundancy. However, raising the modulation order to 64 QAM would bring a 6-Gbps data throughput, while using the same spectrum-occupied bandwidth.

The second approach uses more spectrum bandwidth and hikes data throughput by using a higher symbol rate. For example, the occupied channel bandwidth is approximately 1.22 GHz with a 1-GHz symbol rate , assuming a 0.22 root-raised cosine filter alpha (or excess bandwidth). (A root-raised-cosine filter is frequently used as the transmit and receive filter in digital communications to perform matched filtering and thereby help minimize intersymbol interference.) Increasing the symbol rate by a factor of ten to 10 GHz would raise the QPSK data throughput to 20 Gbps but would use a much wider swath of spectrum (about 12.2 GHz). Raising the modulation order to 64 QAM could increase the data throughput to 60 Gbps. But supporting higherorder modulation schemes at these extreme 6 • 2022

modulation bandwidths becomes much more challenging because of reduced Signal-to-Noise (SNR) ratio, greater linear amplitude and phase impairments, and other technical challenges.

A third approach transmits multiple and independent streams of data for higher data throughput using multiple antenna techniques such as multiple-input/multiple-output (MIMO). However, the actual increase in data throughput would depend on the channel conditions and system overhead.

A review of the first two approaches at H-band (220-330 GHz) from an RF physical layer perspective shows it’s possible to exceed 100 Gbps using 64 QAM modulation with an occupied bandwidth of 30 GHz.

IEEE 802.15.3d The standardization process has yet to begin for 6G. However, IEEE 802.15.3d is an example of an existing standard for fixed point-to-point applications using the sub-THz frequency range between 252 and 325 GHz. In addition, IEEE 802.15.3d defines physical layer modes that enable data rates of up to 100 Gb/sec using bandwidths up to 69.12 GHz. Table 6-17j in the IEEE 802.15.3d specification shows eight different bandwidths supported for the subTHz physical layer.

The bandwidths are multiples of the IEEE IEEE 802.15.3d Supported Bandwidths Bandwidth 2.16 GHz Bandwidth 4.32 GHz Bandwidth 8.64 GHz Bandwidth 12.96 GHz Bandwidth 17.28 GHz Bandwidth 25.92 GHz Bandwidth 51.84 GHz Bandwidth 69.12 GHz

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6 G TES TI NG Conducted Measurement Results at 307 GHz for the 25.92 GHz Bandwidth Test Case with baseband complex pre-corrections applied to the waveform before downloading it to the M8196A AWG.

802.11ad/ay channel bandwidths of 2.16 and 4.32 GHz, respectively.

Row 1 represents the 802.11ad channel bandwidth of 2.16 GHz. Row 2 represents the 802.11ay two-bonded channel (CB2) bandwidth of 4.32 GHz. (Channel bonding in IEEE 802.11 means adjacent channels within a given frequency band are combined to boost throughput between wireless devices.) Row 3 represents the 802.11ay four-bonded channel (CB4) of 8.64 GHz. Rows 4-8 represent integer multiples of the 802.11ay CB2 and CB4 channel bandwidths up to 69.12 GHz. Our sub-THz testbed for H-Band, 220-330 GHz uses a 92 GSa/sec M8196A arbitrary waveform generator (AWG). This AWG generates wide bandwidthmodulated intermediate frequency (IF) signals and has an analog bandwidth of 32 GHz. A compact WR3.4 H-band upconverter converts the IF frequency from the AWG to the desired sub-THz frequency. This upconverter uses a 12x multiplication factor for the local oscillator (LO) frequency.

An E8257D PSG vector signal generator with option UNY provides a low-phase-noise LO for the upconverter and downconverter. A PM5B power meter with WR3.4 waveguide taper measures the power.

On the receive side, the compact WR3.4 H-band downconverter converts the sub-THz frequency to an IF frequency. A UXR multichannel high-performance oscilloscope

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with a sample rate of 256 GSa/ sec digitizes the IF signal. The configuration described here is for an over-the-air (OTA) transmission using diagonal transmitand-receive horn antennas.

Measurements also took place waveguide-to-waveguide using a WR3.4 waveguide bandpass filter, or a WR3.4 waveguide throughsection for wider modulation bandwidth test cases.

This testbed is scalable across D and G sub-THz frequency bands by using different VDI converters and a M8195A 65 GSa/sec AWG. It is also flexible in terms of waveforms because it uses numerous software platforms to generate and analyze candidate waveforms. The testbed supports software written for test applications, as well as system design software and VSA software.

Because the testbed AWG and oscilloscope are multichannel, the number of channels is scalable for MIMO research.

WR3.4 bandpass filter’s bandwidth, so it was removed for this test.

The measured occupied bandwidth is 25.7 GHz, which corresponds to the 21.12 GHz symbol rate multiplied by the root raised-cosine (RRC) filter alpha (21.12 GHz x 1.22= 25.76 GHz).

The symbol rate was 21.12 GHz, so the theoretical raw calculated data rate without forward error correction (FEC) coding rate redundancy comes to 21.12 Gsymbols/sec x 4 bits/symbol (16 QAM) = 84.48 Gb/sec for a single stream of data.

To push the data throughput rate beyond 100 Gb/sec, the 802.15.3d 21.12 GHz symbol rate was increased to 25 GHz. The modulation order was also hiked from 16 to 64 QAM to transmit 6 bits/symbol instead of 4 bits/symbol.

Measurements performed for the OTA test case are shown here. The OTA measurement results include baseband complex precorrections applied and an approximate six-inch edge-to-edge

Measurement examples

References A New Sub-Terahertz Testbed for 6G Research, www.keysight.com/ us/en/assets/7120-1082/whitepapers/A-New-Sub-TerahertzTestbed-for-6G-Research.pdf

Over-the-air (OTA) measurement results at 310 GHz for the 30 GHz bandwidth test case.

Measurements took place for 4.32, 8.64, 12.96, and 17.28 GHz test cases with a WR3.4 bandpass filter and WR3.4 through-section to connect the compact upconverter and downconverter. In addition, for the 25.92 GHz bandwidth test case, the symbol rate was set to six times the 802.11ay CB2 symbol rate (6 x 3.52 GHz = 21.12 GHz).

The channel bandwidth is six times the 802.11ay CB2 channel bandwidth (6 x 4.32 GHz= 25.92 GHz). This bandwidth exceeds the

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spacing between the VDI compact upconverter and downconverter.

The measured occupied bandwidth is 30 GHz, which corresponds to the 25 GHz symbol rate multiplied by the root raisedcosine (RRC) filter alpha (25 GHz x 1.22= 30.5 GHz). The symbol rate was 25 GHz, so the theoretical raw calculated data rate without forward error correction (FEC) coding rate redundancy comes to 25 Gsymbols/sec x 6 bits/symbol (64 QAM) = 150 Gb/sec for a single stream of data. Lowering the modulation order to 16 QAM would yield 100 Gb/sec.

A point to note is this extreme data throughput could evolve into a key performance indicator (KPI) for 6G. IEEE 802.15.3d is an example of an existing fixed pointto-point standard in the sub-THz frequency range between 252 GHz and 325 GHz. Acknowledgment – Keysight Technologies acknowledges Virginia Diodes Inc. (VDI) for providing the VDI H-band WR3.4 220-330 GHz hardware described here.

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Measuring and using static electricity Put away the multimeter if you want to size up static electricity. Here’s why.

David Herres, Contributing Editor

Many

moons ago, a future member of our editorial staff was playing with his very first multimeter. He was a farm kid.

One of his first experiments was to put the negative lead of the meter on a ground rod and the positive lead on one of the metal bands used to reinforce the interlocking concrete staves that formed a feed silo on his family’s farm. To his great surprise, the meter registered a voltage, though the metal band wasn’t connected to anything electrical. Moreover, the higher the metal band on the silo, the greater the voltage.

This kid’s first thought was that his new meter wasn’t working correctly. The mystery wasn’t solved for him until a few years later when college physics instructor explained the concept of atmospheric electricity: Air above the surface of Earth is positively charged, while the Earth’s surface charge is negative. For every meter higher in the air, the voltage rises by around 100 V, i.e. by 100 V/m. This atmospheric potential gradient extends up to about 50 km above the earth. The result is an ion flow from the positively charged atmosphere to the negatively charged surface of the earth.

With this effect in mind, it is easy to see why static discharge is a formidable hazard in (tall) grain elevators due to the possibility of dust explosions. Of course, the atmosphere isn’t the only source of static charges. Electrically neutral objects become charged positively whenever they lose electrons and charged negatively whenever they gain electrons.

In triboelectric charging there is a transfer of electrons from one object to another because of friction.

One classic example of the effect is the rubbing of a vinyl balloon on human hair. The vinyl gains electrons and becomes positive while the hair loses them and picks up a negative charge. But dangerous triboelectric hazards can be present in sawmills or wood-plaining mills in which fine, dry sawdust is suspended in the air. If the dust is blown through a metal duct, the friction of rapidly moving fine, dry wood dust on the inside of the duct can cause static accumulation, electrical discharge and an explosion.

Of course, not all materials exhibit the static electric effect.

Otherwise, we would leave a trail of sparks everywhere we go. Materials that exhibit this interesting phenomenon are known as triboelectric materials, and are members of the triboelectric series. Typically, modern versions of the triboelectric series are based on various properties including the time required for a material to acquire an electric charge relative to other materials in the series and under specified conditions.

The farther apart in the list the two materials appear, the greater the charge transferred. Between metals, there is little or no measurable charge transfer because charge rapidly conducts away from the surface.

Liquids can be triboelectric materials. Fluids characterized by conductivity below 50 pS/m are known as static accumulators because they behave like insulators. Cargo ships and trucks must be as careful when handling static accumulator liquids as when handling flammable liquids: When a static accumulator liquid flows through the pipeline, the pipeline looses some of its electrons. The liquid gains these electrons and becomes negatively charged. When the liquid enters an empty tank, it

An atmospheric electricity demonstrator, otherwise known as a concrete silo.

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ME ASU R ING STAT IC E LECTRI CI TY

An example of a static field meter, the FMX-004.

splashes, again increasing the static charge generation. To avoid this charge generation and potential catastrophes, shippers limit the rate of flow during filling and adopt procedures to reduce the splashing of the liquid in the tank.

Unfortunately it is generally not possible to measure static electricity with a multimeter because the input impedance is too low. Static charge is generally measured in terms of kilovolts and its measurement requires an instrument with a high input impedance.

An electrostatic fieldmeter, also called a static meter, is the usual choice.

It measures the force between the induced charges in a sensor and the charge present on the surface of an object, then converts this force to volts, measuring both the initial peak voltage and the rate at which it falls away.

Measurements usually involve placing a charge monitoring probe close (1 to 5 mm) to the surface of interest while the probe body is driven to the same potential as the measured unknown. Alternatively, electrostatic field meters can use the charge-discharge process of an electrically floating electrode: A corona source charges a floating electrode, which discharges with a regular repetition frequency to the earth-electrode. The discharge repetition frequency is the measured variable which is a function of the background electrostatic field. Beside static charge control in electrostatic discharge (ESD) sensitive environments, another possible application is the measurement of the atmospheric electric field, if sufficient sensitivity is available.

Less destructive and more interesting illustrations of static electricity power can be found on YouTube videos that explain how to construct corona motors powered by atmospheric electricity. Videos we’ve seen use a power source consisting of wire extending about 400 ft high, generally with several sharp points (often sewing

A corona motor powered by atmospheric energy devised by YouTuber RimstarOrg. Visible here are the stationary blades and the connections to the wire in the sky.

pins) at the top to concentrate charge.

The setup generates a current in the low microamp range.

The corona motor powered by the atmospheric energy is a form of electrostatic motor containing an even number of stationary blades surrounding a cylinder which rotates. The cylinder is an electrically insulating material-plastic soda bottles are a popular choice.

Electrically conductive material such as aluminum foil lines the inside of the cylinder. The blades are electrodes. They are charged such that one is negative, the next positive, the next negative, and so on. Thus every other blade connects to the wire in the sky, and the alternate blades connect to ground.

In operation, a negative electrode sprays negative charge onto the section of the cylinder it faces. Like charges repel each other so the negative charge on the electrode repels the negative charge on the cylinder. But charges can’t go anywhere on the electrical insulator cylinder. Instead they repel, pushing the cylinder, and the cylinder begins to rotate. The blades are usually at an angle to the cylinder, causing charge to spray on the cylinder more in the direction angle.

As it rotates, the negatively charged section of the cylinder approaches the next electrode which is positively charged. Unlike charges attract each other. So the negatively charged section is attracted by the positively charged electrode. As the negatively charged section passes the positively charged electrode, ionization removes the negative charge from that section, leaving it positively charged.

The now positively charged section is repelled by the positively charged electrode. The positively charged section approaches the next electrode which is negatively charged. The unlike charges attract, and the positive section is attracted by the negative electrode.

Common triboelectric materials, from most positive to most negative Nylon Glass Acrylic Quartz Lead Cat's fur Silk Aluminum Amber Resins Rubber Copper Brass Silver Gold Platinum Polyethyline PVC Silicon Common triboelectric materials.

References Simco Ion, www.simco-ion.co.uk/ products/static-measurement-instruments/ How Powering with Atmospheric Electricity Works, https://www.youtube. com/watch?v=2rVdEhyMR6A

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Key considerations for RF power measurement equipment

Bob Buxton, Boonton Electronics

The choice of power sensor and meter greatly depends on the qualities of the signal being measured.

Thermal and detector type measurements are “direct sensing,” in which the amplitude of the RF signal applied to a load element is measured by converting the RF to an easily measured dc quantity.

Today’s

In a USB-connected sensor, temperature compensation takes place in the sensor. The digital connection is unaffected by temperature variations, so a calibrator is not required. A USB sensor may be used with a power meter for a dedicated benchtop experience, or with a PC and analysis software.

When settling on a technology for RF power measurement, there are a few basic questions that require answers.

• •

complex modulation schemes and pulsed

communication modes have brought the need to measure RF power ever more accurately and efficiently. There are multiple technologies available for the measurement of RF power,

such as pulse rise/fall times, pulse width, pulse repetition interval (PRI), and more. Power meter/sensor combinations may be classed as CW, average, or peak. Displays may be numerical or graphical, displaying the powertime profile of a signal.

Traditional benchtop systems employ a sensor that provides an analog signal to the

• • •

What are the minimum and maximum carrier frequencies?

What is the minimum and maximum expected power level (i.e. dynamic range)?

Are connections coaxial or via a waveguide?

Is the characteristic impedance 50 Ω, or something else?

Is the carrier CW or modulated via pulse, analog, or digital modulation?

falling into four categories: Thermal: Measuring the heating effect of RF power upon a sensing element.

Diode detector: The RF signal is rectified or “detected” to yield a dc voltage proportional to the signal amplitude.

Receiver: A tuned circuit receives the signal to allow measurement of its amplitude component.

RF sampling: The RF signal is treated as a baseband ac signal and is directly digitized.

Thermal and detector type measurements are “direct sensing,” where the amplitude of the RF signal applied to a load element is converted to an easily measured dc quantity.

This RF-to-dc conversion typically takes place close to the signal source via a small converter probe known as an RF power sensor that connects to the device under test (DUT).

We will consider in detail detector-based direct sensing. A power meter may display a variety of results such as average or peak power and automatically measure parameters

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Typical power meter display showing average, minimum, and maximum readings power meter. But today there are also USBconnected sensors where all detection and digitization takes place in the power sensor which transmits digitized data to the power meter over USB. Traditional analog-connected systems require calibration because the analog signals can vary slightly with temperature.

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Similarly, is the video bandwidth narrowband or wideband?

If the signal is pulsed, how fast does it rise?

Next, consider the signal measurements that might be required: Just average levels? Peak eeworldonline.com

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R F M E ASUREMENTS

PMX40 power meter and USB sensors showing power-time profile.

information (peak-to-average power ratio)? Time-domain measurements (pulse profiling and automated pulse measurements)? Will there be a statistical power analysis?

Decisions that may be less obvious arise from the type of signal being measured: CW, modulated, or pulsed. Several questions in this area arise frequently from users: Q: I want to measure average power. What type of sensor should I use and how would I know that I

Factor = PEP/Average Power) or only during the pulse (Crest Factor = PEP/Pulse Average Power).

of pulse power measurement.

What do I need to know about this measurement?

A: Envelope power is the amplitude change due to modulation or distortion as a function of time, averaged over one or a few cycles of the RF carrier signal. Peak envelope power (PEP) is just the highest peak amplitude

also make a series of automated pulse measurements such as rise/fall time, overshoot, droop period, duty cycle, off-time, duty cycle, and more. Some can make measurements on multiple pulses in a burst.

Q: I need to measure a pulsed signal and the power sensor datasheet mentions “video bandwidth,” “continuous sample rate,” and “effective sample rate.” What’s the relevance of these parameters?

Measurement of multiple pulses in an IFF/secondary surveillance radar squawk burst

To make these measurements it is necessary to use a peak power sensor with a peak power meter or a USB-connected peak power sensor with either a meter or analysis software running on a computer. Some systems can

A: When measuring pulse signals there are three particularly important things to consider: pulse rise time, minimum pulse width to be measured, and the time resolution required.

Rise time is determined by the

The distinctions between envelope power (blue), peak envelope power (PEP) (green), average power (red), and pulse average power (black). have sufficient video bandwidth for my measurement?

A: For average power measurements a suitable sensor would be a thermal or a realtime true average power sensor.

Because the measurement result is an average over time and there’s no need to track the signal profile, the amount of video bandwidth is unimportant. Sensors are available that can deliver 100,000 measurements per second.

Q: I need to make pulse power measurements, but I have heard that there are multiple types eeworldonline.com

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due to modulation. Pulse average power is the average power level during the pulse. And the average power is the total power over the pulse repetition interval, which includes both the signal burst and the time interval leading up to the next pulse.

Using these distinctions, other derived measurements can be determined as well (principally, crest factor). Crest factor is defined as peak-to-average-ratio (PAPR). From this definition, crest factor can be the ratio of the PEP to either the average power over the PRI (Crest

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In this example of RIS, the continuous sample rate is 1/(time between samples for each acquisition). A profile of the waveform is built up by multiple sample acquisitions which vary in delay time from the trigger point. An internal clock running asynchronously to the trigger signal ensures that the samples are taken randomly with respect to the trigger. Samples from each acquisition are interleaved between those from previous acquisitions.

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RIS power measurements (left) with continuous sampling of 100 MSa/sec and a time-base resolution of 100 psec uncover artifacts not visible when making conventional measurements (right) even when displayed at the same time-base setting (10 nsec/div) video bandwidth (VBW). Rise time ≈ 0.35/VBW. A sensor with 195 MHz VBW would be suitable for rise/fall time of about 2 nsec. However, such a sensor would probably be specified at the more conservative rise time of less than 3 nsec.

The terms “continuous” and “effective” sample rate are relevant for sensors that use Random Interleaved Sampling (RIS), also called Repetitive Random Sampling (RRS). In equivalent-time sampling, a picture of a waveform is created over time by using a series of samples taken from repetitive waveforms. RIS is a form of equivalent-time sampling that boosts the apparent sample rates of repetitive signals by combining several triggered waveforms. Because the trigger time happens randomly for each sample acquisition, the digitizer samples different points in the waveform on consecutive acquisitions. By combining these waveforms, RIS sample rates can be up to 100 times higher than the ADC sample rate on the digitizer.

Power sensor/meter combinations and USB power sensors are available that have sample rates up to 100 MSa/sec. The minimum pulse width is determined by the continuous sample rate. To ensure that at

least one sample coincides with the pulse, the pulse width cannot be less than the time between samples. For a continuous sample rate of 100 MSa/sec, that corresponds to a minimum pulse width of 10 nsec.

The effective sample rate is determined by the resolution of the random delay from the trigger point. For a sensor with an effective sample rate of 10 GSa/sec this means a signal may be measured with a time-base resolution of 100 psec. Figure 6 shows an example.

Q: Can I assess the linearity of my amplifier using a power sensor and meter?

A: Yes. The key is to select a power meter/sensor combination or a USB-connected sensor that can make statistical measurements such as crest factor and complementary cumulative distribution function (CCDF).

Performing these measurements on the amplifier input and output signals makes it possible to see at what point distortion starts. Boonton’s Testing High Power GaN Amplifiers for Radar Signals using Peak Power Meters contains more information on statistical measurements.

In summary, CW, average, and peak power sensors and meters all have their place. However, if multiple types of power measurement are needed and budgets allow only one type, the peak power sensor and meter may be the best choice because it can make all three major types of measurement.

References Principles of Power Measurement, A Reference Guide on RF & Microwave Power Measurement. https://boonton.com/resource-library/principlesof-power-measurement-guide.

What Do You Want to Measure – Peak or Average? https://boonton.com/Portals/0/docs/boontonprinciples-of-power-measurement-3-what-do-youwant-to-measure-peak-or-average.pdf

Left – CCDF curve of a modulated signal (yellow amplifier input, green amplifier output)

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Testing High Power GaN Amplifiers for Radar Signals using Peak Power Meter. https://boonton. com/applications/communications/artmid/855/ articleid/86/testing-high-power-gan-amplifiers-forradar-signals-using-peak-power-meters

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TE ST & M E AS U R E M E N T HA N DB OOK A compact lab bench setup with prototype PCB being tested using Moku:Go.

Test and measurement in education Lab classes become much more meaningful when students can operate their own test instruments rather than watch an instructor turn knobs and push buttons.

Paul Cracknell, Liquid Instruments

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Consider

the typical EE lab in the average university. Labs must be outfitted with costly equipment to give students exposure to the breadth

of measurements that characterize electrical engineering and related disciplines. When there aren’t enough instruments for individual, hands-on experience, students must rely on shared, group demonstrations or simulation.

The past couple years have taught many of us in engineering fields that we can indeed be productive working remotely. But ultimately, we often need in-person work and real hardware to get the job done. Ditto for teaching institutions. Even the best video conference cannot replace hands-on interaction and in-person lab classes.

In 2021, Liquid Instruments introduced Moku:Go as an integrated benchtop hardware platform, small enough to take home if needed. It integrates a suite of instruments that would be present on any undergraduate lab bench (an oscilloscope, waveform generator, programmable power supplies) as well as those that would typically be less accessible (spectrum analyzer, frequency response analyzer, lock-in amplifier, PID controller, and FIR and IIR filtering tools).

During the early days of the COVID-19 pandemic, we at Liquid Instruments were faced with a unique opportunity to help remote labs by using the multi-instrument Moku:Go. We saw that as much as students benefited greatly from being able to take a Moku:Go home and run experiments and lab exercises, this fell short of expectations of most engineering undergraduates from the perspective of interpersonal collaboration. As much as engineering is a practical discipline, it is a collaborative one as well; working with lab partners on a problem, discussing with fellow students, and having the presence of an

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ED UCATI O N

Students using laptop UI with Moku:Go to perform a power management lab. experienced lab assistant ready to offer guidance cannot be as easily replicated remotely.

While virtual tutoring sessions and UI screen sharing were effective short-term solutions, Liquid Instruments was fortunate to host a number of COVID-19 safe student lab sessions, some outdoors in the fresh air, others with appropriate social distancing measures. These were held at several colleges including UCSD, Penn State, and the University of Arkansas.

Accessing multiple instruments in a single, portable device enabled enhanced flexibility in a world that required it.

Today’s students have grown up with smartphones and tablets with modern user interfaces with touch and gesture controls that are user-centric and task-oriented. A physical interface of a traditional oscilloscope or waveform generator presents a large array of buttons and dials and a deep menu system.

This style of interface has long since vanished from consumer products and as such presents a barrier to learning. The student spends more time working on the oscilloscope interface than learning the subject matter, be it an op-amp filter or a PID controller.

Decisions about the best test and measurement equipment for the classroom must now consider usability and learnability,

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as these can greatly enhance the students’ experience and confidence. An overly complex UI can be intimidating and detracts significantly from subject matter learning. Teaching assistants at the undergraduate level consistently report that up to half of their valuable lab time goes toward troubleshooting test equipment issues. The primary focus should be on effectively teaching the technical subject at hand and not trying to get the equipment to work, which is frustrating for the department and student alike.

Integrated test combined with an intuitive user interface allows universities to simultaneously upgrade their lab equipment and more effectively go from simulation to practice. The University of Detroit Mercy implemented a variety of labs across the EE curriculum using Moku:Go, including standard foundational lab work such as the investigation and understanding of a Texas Instruments buck regulator and an introduction to the basics of opamp filter design.

Power supplies are a key element of any lab bench.

Integrating them into Moku:Go alleviates the need for a separate bench supply. The student can then focus on using the probes of the oscilloscope and connections of the swept sine waveform generator.

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To give students a sense for the impact of the filter, the lab exercise starts with students stepping the waveform generator through a frequency range and taking manual, step-by-step measurements of the output amplitude and phase across frequency. This real hands-on lab experience enables students to visualize the signals and explore various operating points.

After manually plotting perhaps a half dozen points, students can naturally progress to using the integrated Frequency Response Analyzer (FRA) instrument to rapidly/automatically plot the Bode chart. With this sequence, analog, op-amp based filters can be covered in two or three labs so students can move to the digital or DSP-based filter solutions which now dominate modern designs.

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Bridging simulation and implementation Simulation is critical for electrical engineering design, but transitioning to a real hardware implementation is the ultimate goal. It’s crucial to help students understand how to make that transition and correlate results between simulation and physical measurements. MATLAB can help relate the mathematical foundation of filters to real-world responses.

For example, students can use MATLAB filter designer to design a specific filter response.

When the MATLAB simulation has produced a desired response, Top, Moku oscilloscope and waveform generator UI. Below, intuitive frequency response analyzer showing amplitude and phase vs. frequency.

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TE ST & M E AS U R E M E N T HA N DB OOK the filter coefficients can then be exported and uploaded to Moku:Go’s digital filter tools to realize the filter in hardware. Once the filter is running on real hardware, an oscilloscope and waveform generator, or FRA can then be used to confirm the frequency response.

This ‘closes the loop’ on the design process and gives students a complete view of a design cycle. Together with a modern UI on a multi-functional instrument, this makes for an enjoyable and rewarding student experience while focusing attention on the subject matter rather than on the tools themselves.

Proportional-integral-derivative (PID) controllers find use in numerous engineering settings. No wonder, then, that their mathematical and theoretical underpinnings are extensively covered in textbooks and classroom lectures. But a solid lab tutorial is invaluable for grasping the use of PID controllers to the real world. That’s why we designed a PID controller instrument for Moku:Go. This capability allows a lab to be readily written around the PID controller where students can adjust parameters in real-time and see the impact on the loop response.

Our focus has been primarily on ensuring Moku:Go addresses the needs and learning expectations of undergraduate. But it is also important to consider budget constraints and logistics of educational institutions. A typical undergraduate lab might be budgeted for a number of stations containing power supplies, oscilloscopes and waveform generators. If the lab contains a spectrum analyzer, PID controller or Lock-in Amplifier, there’s generally only one or two; so only larger groups can use them.

With Moku:Go’s integrated instrument suite, personalized work can replace larger group-based learning experiences without straining department budgets. Software-defined FPGA instrumentation like Moku:Go places a Lock-in Amplifier in the hands of students so they can perform AM and FM demodulation on their own, something difficult to accomplish with the makeup of legacy teaching labs.

Moku:Go also aims at a higher goal of equalization of learning outcomes. It is wellknown that students learn at different paces.

So it can be challenging to get a whole class on equal footing at the end of each school year. This is doubly true in engineering where lab time is limited. Portability and integration in hardware make things easier for students who need extra time. We want every student to graduate as the best engineer they can be,

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Moku:GO PID interface. and that means providing them with as much hands-on experience in the lab as possible.

We learned a lot about engineering and science education during the COVID-19 pandemic. While remote, video conferencing works to a certain extent for lectures, it is a poor substitute for in-person, hands-on laboratory learning. Engineering is by its nature theoretical and mathematical, but ultimately practical. Every graduate engineer must translate theoretical knowledge into real-world practice and solutions for society.

Practical lab experience is critical in providing solutions to real world problems.

Today’s undergraduates have high expectations of user interface for the devices they use in their everyday life. The next generation expects their test and measurement equipment to live up to these standards - standards that allow them to do their best work.

References Liquid Instruments, https://www. liquidinstruments.com/education MATLAB filter designer (https://www. mathworks.com/help/signal/ug/introductionto-filter-designer.html)

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MIT I G ATI NG EMI

EMI testing for IoT transceivers Leland Teschler, Executive Editor

A few oscilloscope measurements can give insight into sources of electromagnetic interference that garble the operation of wireless IoT devices.

Specialized instrumentation is available for testing electronic systems against EMI and radiated emissions standards. But oscilloscopes are generally the goto instrument for isolating problem areas while IoT gear is in the development phase. A few tips can help speed the debugging process when scopes are involved.

Many of the interference problems associated with IoT wireless devices manifest as noise that prevents the IoT

simultaneously. For a cellular IoT device, TIS is a measure of how far away the device can be from a cell tower before the cell signal is too weak to receive. Thus TIS is a receive/downlink test.

(Note that there is also a transmit/uplink test called TRP (total radiated power) that adds up all the power the device transmits at a particular channel of a cellular transmit band. The better the TRP, the further from a cell tower the device’s signal can

Higher end oscilloscopes often have facilities that allow time and frequency-correlated displays such that the frequency components corresponding to specific parts of the time display can be isolated.

This example is from a Tektronix scope which is displaying the timebased signal and frequency components generated by a car FOB.

Examine

a typical

IoT electronics, so sensitive RF

wireless

receivers often sit nearby other

device with an internet of things

circuitry that can potentially

connection and you’ll likely find

interfere with their operation.

multiple RF receivers squeezed

This reality has put a premium

together in close proximity

on being able to detect and

onto a circuit board. Cramped

correct interference problems on

quarters characterize modern

embedded systems.

Currents and voltages in a buck converter

An analysis of a buck converter showing the main loops of current flow. The dotted line denotes a loop having the highest di/dt.

Takeshi Murase at the KTH School of Electrical Engineering and Computer Science performed the analysis. eeworldonline.com

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A waveform from a buck converter where ringing is visible. Ringing frequencies and their harmonics can be large contributors to converter EMI. Takeshi Murase at the KTH School of Electrical Engineering and Computer Science performed the analysis. device from passing TIS (total isotropic sensitivity) tests dictated by cellular carriers. TIS is the minimum power level of a cellular signal the IoT device can reliably receive via its cellular antennas in all directions. For low-throughput cellular technologies like LTECAT-M1 and NB-IOT, transmit and receive functions share one antenna. A cellular product with multiple antennas can undergo TIS testing for each antenna separately or for both of them

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be received. But TRP performance is generally unaffected by EMI the IoT device generates.)

IoT devices operating in the U.S. must have TRP and TIS figures that exceed minimum levels set by cellular carriers in the U.S. Otherwise the IoT device can’t be certified for use on any cellular carrier whose minimum requirements it doesn’t meet.

Cellular TIS is measured on a single channel within a cellular downlink band. Three receive

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Makers of converter chips take several precautions to minimize the EMI their ICs generate.

For example, note the PCB layout for this switcher chip, the LT8640S, which features supertight component spacing to minimize radiating loops. Additionally, the chip employs spread spectrum frequency modulation to spread radiated energy over a wide spectrum as a means of reducing the energy in any particular part of the spectrum. channels are generally tested per band. The exception is LTE Band 13 used in the U.S. primarily by Verizon which is only tested at one receive channel centered at 751 MHz. The minimum allowable TIS for certification is generally around -99 dBm. Problem is, high-frequency switching or digital signals can interact with nearby conductors, including those residing within the

PCB, to produce electromagnetic radiation.

If that radiation includes frequencies (spectral content) in the cellular receive bands, the radiation can drown out low-level cellular signals in those bands, thus compromising TIS test results and the IoT device’s ability to receive cellular signals from cell towers. Often, EMI is said to “desense” receivers when it is high enough to have an impact on performance.

An old trick for pinning down noise sources: Disable the dc supply and substitute a battery while looking for changes in EMI.

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EMI design reviews of IoT devices tend to focus on bus noise, clock signals, and particularly dc-dc converters as the main culprits contributing to noise energy. Noise sources couple into nearby circuits via four typical paths: radiation from cables, through common impedance coupling where a noisy source shares the same current return path as a noise-sensitive circuit; and through capacitive and inductive coupling.

Capacitive coupling arises where parallel conductors essentially form the plates of a capacitor. The classic example on a circuit board is the capacitance between a heat sink and a PCB return path or trace. The capacitance allows signals with a high dv/dt to pass. Similarly, inductive coupling arises between parallel wires that induce current in nearby conductors. The inductance allows signals with a high di/dt to pass.

The types of signals that get inductively or capacitively coupled include clock signals.

Clocks tend to generate narrow-band EMI consisting of spikes located at the clock frequency harmonics.

In contrast, dc-dc converter noise tends to be broad band and manifests itself as a rise in the noise floor of the receiver. The broadband noise is not because of the converter’s fundamental switching frequency, which generally is in the 3 MHz range or below.

Rather, noise tends to arise because of the fast rise time of the converter switches.

Rise times can be on the order of a nanosecond. Worse, there can be ringing on the switching waveform. Given that the ringing happens after the rise or fall time of the converter switch, the ringing waveform rise and fall times have a more severe impact on EMI. There are peaks in the emissions spectra corresponding to the ringing frequency and its harmonics. All in all, fast power switching and ringing waveforms from

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MIT I G ATI NG EMI Diagnosing problems

Near-field probes can be had for a variety of frequency ranges, but a do-it-yourself approach with a simple coil of wire on the scope probe also works. The main difference between these approaches is the amplitude of the signals captured; rise times, and other parameters are the same.

There is also at least one YouTube video on how to make a DIY near-field probe. today’s converters can generate EMI well above 1.5 GHz. PCB layout practices can be another source of EMI. Experts says power and signalreturn PCB layers should be spaced no further than 3 mils apart, a practice frequently ignored in four-layer boards. Another suggestion: Put the ground return plane on the outer PCB layers so they can be stitched around the edge of the board to facilitate the construction of a Faraday cage. For similar reasons, PCB layouts of dc-dc converters should minimize the board area encompassed by conductor loops where there is a high di/dt.

One of the main tools for discerning radiated emissions is a near-field H probe. Scope makers as well as other manufacturers make these devices which often come in kits for different frequency ranges, typically spanning from about 10 kHz to 3 GHz. They are basically inductive loops. Holding the loop near the radiator allows it to pick up radiated noise by induction. The waveform captured by inductive probes have the same rise time, ringing, pulse period, and other qualities as waveforms captured directly with a conventional scope probe. The only difference is the amplitude of the captured signal.

One advantage to using a near-field probe is that there’s no need for direct access to a circuit probe point. Additionally, nearfield probes avoid the possibility of shorting a power circuit with a scope probe when power components are densely packed. The disadvantage of the near-field probe is that it may only get close to a radiating source but may not be able to precisely locate it.

A way of getting a fix on the radiating power supply component is to disconnect the supply and temporarily substitute a battery, noting whether or not the noise disappeared.

In the same vein, converters often contain a main power inductor that can be disconnected, which disables the switching action.

Also helpful is a facility found on many higher-end scopes is time and frequency correlation. This capability allows the scope to trigger on an event and simultaneously generate both the voltage/time display as well as the frequency display so that frequency components corresponding to specific parts of a zoomed-in

waveform time display can be examined.

Another helpful component is the RF current probe, useful for detecting harmonic currents on cables. These resemble ordinary clamp-on current probes but handle upper frequencies in the 200 to 500 MHz range and can detect harmonic currents low enough to be measured in microamps. These probes are basically current transformers where the cable under test is the primary and the probe is the

A handy device for gauging cable harmonic currents is an RF probe. This model is the EZ-17 probe from Rohde&Schwarz which works up to 200 MHz. secondary, loaded by the scope 50-ohm input.

RF current probes on cables are for measuring conducted rather than radiated emissions. They are useful for ensuring devices with cables pass the FCC Class B radiated emission test which permits 6 to 8 µV radiated fields at most.

References Analog Devices silent switcher, https:// www.analog.com/media/en/technicaldocumentation/user-guides/DC2530AF.PDF TekBox TEM cell, https://www.tekbox.com/ product/Tekbox_TEM_Cell_Manual.pdf Tektronix, https://www.tek.com/en/blog/3important-insights-from-simultaneousfrequency-and-time-analysis Rohde&Schwarz, www.rohde-schwarz.com

A TEM cell is basically a desktop anechoic chamber for testing of radiated emissions.

Tekbox developed open TEM cells to cover the frequency range up to 2 GHz and beyond. Combined with a spectrum analyzer, products can be tested before and after EMC related design modifications. eeworldonline.com

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What you need to know about error analysis in PCIe 6.0 Designs The much-awaited next generation of PCIe makes it important to understand bit error-rate measurements.

Hiroshi Goto, Anritsu Co.

PCI

Express (PCIe) 6.0 is being developed to meet the high-

speed data transmission needs of emerging applications, particularly data centers supporting 5G. It features a doubling of data rates and other enhanced performance specs but at a cost of added complexity for History of PCIe data rates, courtesy of PCI-SIG.

high-speed interconnect designs. Engineers designing such equipment must verify performance via real-time analysis, an approach that saves time and improves repeatability.

To accommodate faster designs, PCIe 6.0 utilizes 32 Gbaud PAM4 signaling. (Basically, PAM4 is a modulation scheme that combines two bits into a single symbol with four amplitude levels. This effectively doubles a network’s data rate compared to that for 1/0 high-low signaling.) The first five generations of the specification all used Non-Return to Zero, NRZ. (As a quick review, Return to Zero signal transmission of a logic “1” will always begin at zero and end at zero whereas NRZ signal transmission of a logic “1” may or may not begin at zero and end at zero.) PAM4 allows the specification’s channel reach to remain similar to that of the PCIe 5.0 specification. As is the case with all previous PCIe generations, PCIe 6.0 is fully backwards compatible, so NRZ will also be supported.

Though the PAM4 scheme doubles the transfer speed, this approach reduces the bandwidth per bit and degrades the waveform, thereby creating a small eye. In other words, the additional signal states of PAM4 signals make them more susceptible to errors than NRZ signals. The underlying frequency is the same as the PCIe 5.0 specification at 32.0 GT/ sec NRZ, but there is extra circuitry and logic involved for the PAM4 mode in PCIe 6.0. This

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is necessary to track three eyes, along with the logic changes needed to operate in what’s called Flow Control Unit (FLIT) mode.

The PCIe 6.0 specification introduces FLIT encoding. FLIT encoding takes place at the logical level to break up data into fixed-size packets. To quickly review, a FLIT is a logical unit of information. A network packet is composed of FLITs. The first FLIT in a packet is the header FLIT and holds information about the packet’s destination address. Subsequent body FLITs contain the actual data payload, and the final

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tail FLIT performs book keeping to close the connection between the two nodes.

Previous versions the PCIe spec employed no forward error correction. But defining the logical layer in fixed-size packets enables PCIe 6.0 to implement FEC and other error correction methods because such methods require fixed-size packets. Once the link A comparison illustrating the difference between the bit patterns and eye diagrams for NRZ and PAM4.

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PCl e 6.0

The evolution of the Ethernet standard, courtesy of the 802.3 Beyond 400 G Study group. 400 GbE is in use for current Ethernet designs, with next-generation products and systems under development utilizing 800 GbE and even 1.6 TbE. The integration of 400 GbE/800 GbE into emerging high-speed designs makes performance verification more challenging. For example, signal-tonoise (SNR) measurements are more complex when conducting 400/800-GbE measurements. operates in FLIT mode, any speed change to lower data rates will also have to use the same FLIT mode. Once enabled, FLIT mode is followed in the link, regardless of the speed. The improved bandwidth that results from low overhead amortization allows for high bandwidth efficiency, low latency and reduced area.

PAM4 signaling experiences less channel loss because it runs at half the frequency with two bits per unit interval, UI, compared to 1/0 signaling. One byproduct, however is that there is a 10-dB SNR reduction. The three eyes associated with PCIe 6.0 are in the same UI. The result is a reduced eye height and width. Consequently, the bit error rate (BER) is several levels of magnitude higher with PAM4.

For PCIe 6.0, BER is a combination of the First Bit Error Rate (FBER), correlation of errors in a lane, and correlation of errors across lanes. FBER is the probability of the first bit error happening at a receiver in a link.

The PSI-SIG conducted extensive studies before determining that the FBER in PCIe 6.0 is 10-6.

PCIe 6.0 uses a unique approach to maintain low latency for these high-speed applications.

It integrates lower FBER with lowlatency Forward Error Correction (FEC) for initial correction. FEC is an advanced coding technique that transmits the necessary data to correct errors through the PAM4 link. It serves as a key technology to assure transmission quality. It is an essential element in testing because of the reduction in SNR caused by PAM4. eeworldonline.com

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Once FEC takes place, a robust cyclic redundancy check (CRC) detects any errors that remain. The result is a link-level retry mechanism to ensure PCIe 6.0 meets low latency, high-bandwidth, and high-reliability requirements.

The PCI-SIG has established a low latency FEC of below 2 nsec for PCIe 6.0, and that is to be part of the specified overall signal latency of below 10 nsec.

FEC is based on a fixed number of symbols. Consequently, it is simple to transition to FLITs, as they are of a fixed size as well.

Evaluating performance A recommended approach for evaluating performance is to establish an FEC symbol-error threshold. Use of a threshold gives engineers broader control over error conditions that affect patterns during capture by ignoring insignificant events that are normally corrected in the FEC environment.

To set a threshold, a BERT generates a PAM4 signal to the device under test (DUT) receiver input. The DUT determines the logic state of the input signal and loops its decision to the transmitter output. A BERT’s error detector (ED) determines if the DUT’s decision was correct. Here, the BERT’s jitter and noise profiles must comply with standards. And the BERT used to conduct the FEC symbol error measurements should have a high-sensitivity 116Gbit/sec PAM4 ED. When conducting the test, it’s important to note that a random error is not as meaningful as one happening in a burst. Also

designworldonline.com

important: Some burst errors cannot be corrected by FEC beyond a certain limit. Post processing must take place beyond that limit to help determine why the DUT might be misreading an incoming symbol.

With this approach, engineers can evaluate a device using standard PRBS (pseudo-random binary sequence) patterns while basing error detection capture on events that might be problematic in an FEC environment.

If the input data is captured once the number of FEC symbol errors exceeds the threshold setting, an FEC symbol capture measurement should take place to determine which data stream causes the uncorrectable errors. An uncorrectable burst error is defined as Reed-Solomon (RS) FEC symbol errors that total more than 16 per code word.

A BERT test solution with real-time FEC symbol-capture capability makes for repeatable and

high-confidence measurements.

Engineers can monitor changes in bit errors and FEC symbol errors with deviations in input amplitude and jitter conditions in as they arise.

The input data is captured when the number of FEC symbol errors exceeds the threshold setting, up to 128 burst error events. The causes of FEC-uncorrectable errors can be analyzed from the captured data more efficiently as a result.

The integration of PAM4 technology into PCIe 6.0 has allowed next-generation interconnects to meet the requirements of emerging high-speed applications. A comprehensive test solution that includes real-time FEC analysis capabilities helps engineers verify designs and have greater confidence in product performance.

References Anritsu Co., www.anritsu.com

A test solution with real-time FEC symbol capture capability provides multiple advantages in high-speed design verification.

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